Non linear controlled power supply with a linear soft start operation

ABSTRACT

A system for controlling a soft start for a switching converter includes a digital feedback control system for receiving an analog voltage signal representing the output of a switching converter, for digitally processing the analog voltage signal by comparing it to a corrected reference voltage to generate a voltage error signal and for generating drive signals to control the operation of the switching converter to provide a regulated output. A microcontroller provides a secondary feedback loop for generating the corrected reference voltage. The secondary feedback loop enabling the output voltage signal to vary linearly over time during a soft start.

CROSS-REFERENCE TO RELATED APPLICATIONS

This is a Continuation-in-Part of U.S. Ser. No. 11/172,484, filed onJun. 30, 2005 entitled “DIGITAL POWER SUPPLY WITH PROGRAMMABLE SOFTSTART” filed Jun. 30, 2005, which is a Continuation-in-part applicationof U.S. Ser. No. 11/096,597, filed on Mar. 31, 2005, entitled “DIGITALPWM CONTROLLER” and it is related to U.S. Pat. No. 7,245,512, andentitled “PID BASED CONTROLLER FOR DC-DC CONVERTER WITH POST-PROCESSINGFILTERS” issued Jul. 17, 2007 and U.S. patent application Ser. No.11/096,853, filed Mar. 31, 2005, and entitled “DIGITAL POWER SUPPLYCONTROLLER WITH VOLTAGE POSITIONING” each of which are incorporatedherein by reference in their entirety and claims priority in ProvisionalApplication No. 60/591,463, filed Jul. 27, 2004, which is incorporatedherein by reference in its entirety.

TECHNICAL FIELD OF THE INVENTION

The present invention relates to power supplies, and more particularly,to power supplies having a linear soft start operation.

BACKGROUND OF THE INVENTION

DC-DC power converters are utilized in situations where one DC voltageis converted to another DC voltage. In one application, that associatedwith PC based systems, the processor requires a fairly low voltage and afairly high current. Rather than convert an incoming AC voltage down toa very low DC voltage and then route the low DC voltage across a PCboard, a higher DC voltage is output by the power supply, routed aroundto the various components on the PC board and then, proximate to theprocessor, the voltage is down converted to a very low level on theorder of 1.0 V. This requires a conversion device to be disposedproximate to one or more high current integrated circuits on the board.

Typical DC-DC converters are fabricated using a switching supply thatutilizes a switched inductor or capacitor configuration with the inputDC voltage switched to the input thereof with a periodically waveformoperating at a preset switching frequency with a varying duty cycle. Bysensing the output voltage and comparing it with a desired voltage, theduty cycle of the waveform can be adjusted to control the amount ofcurrent supplied to the reactive components. This control is facilitatedwith a negative feedback control loop.

There are two types of feedback loops, an analog feedback loop and adigital feedback loop. The analog feedback loop is well understood andprovides some advantages over the other type of feedback loop, thedigital feedback loop. Each of the feedback loops has associatedtherewith a voltage sense input for sensing the supply output voltageand a pulse width modulator (PWM) for generating switching pulses fordriving switches. The sensed voltage is compared in the analog domain toa desired operating DC voltage to generate an error voltage that isreduced to essentially zero volts at regulation. To compensate for loopphase shift, there is provided a compensator. This provides some phaselead in the feedback loop for the purpose of loop stability. The digitalcontroller portion of the digital feedback loop is similar to the analogfeedback loop. The voltage signal sense input utilizes an analog todigital converter (ADC) to convert the output voltage to a digital valueand then compare this to a desired voltage to determine the differencevoltage as an error voltage. A digital compensator then provides somephase lead to the feedback to maintain stability in the control loop.This digital error voltage is then converted into a varying pulse widthfor output to the driving switches on the switching converter. This ineffect is a digital to analog converter. Typical switching converterssuch as buck converters can utilize single or multiple phases tofacilitate the switching operation.

During soft start the input voltage of the power supply is controlledfrom “0” to the reset point of the power supply. With analog controllersa capacitor is used to control the slope of a line between the “0” pointand the reset point. With existing analog controllers only the slope ofthe line may be charged. It would be desirable to have some manner formore particularly controlling the input voltage during a soft start overthat presently provided by analog controllers.

Another problem with some digital power supply controllers is theinability to provide linear operation over all voltage ranges. While thepower supply will operate linearly within higher voltage ranges of thepower supply, during for example, a soft start process, the power supplyhas nonlinear operation within the lower voltage ranges. Some manner ofenabling the power supply to operate linearly over all voltage rangeswould be particularly useful in certain applications.

SUMMARY OF THE INVENTION

The present invention disclosed and claimed herein, in one aspectthereof, comprises a system for controlling a switching converter. Thesystem includes a digital feedback control system that receives ananalog voltage signal representing the output of the switchingconverter. The digital feedback system digitally processes the analogvoltage by comparing it to a corrected reference voltage to generate avoltage error signal. The voltage error signal is used to generate drivesignals to control the operation of the switching converter to provide aregulated output. A microcontroller provides a secondary feedback loopfor generating the corrected reference voltage signal. The secondaryfeedback loop enables the output voltage signal to operate linearly overlow voltage ranges.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention and theadvantages thereof, reference is now made to the following descriptiontaken in conjunction with the accompanying Drawings in which:

FIG. 1 illustrates an overall block diagram of a switching power supply;

FIG. 2 illustrates a schematic diagram of the switching portion of ahalf-bridge power supply;

FIG. 3 illustrates the timing diagram for the control pulses to theswitching power supply;

FIG. 4 illustrates a diagrammatic view of the digital controllerutilized in conjunction with a buck converter;

FIG. 5 illustrates a more detailed view of the digital controller;

FIGS. 6 a and 6 b illustrate a block diagram of the microcontrollerportion of the digital controller;

FIG. 6 c illustrates a diagrammatic view of a monolithic solutionutilizing the embodiments of FIGS. 4, 5, 6 a and 6 b;

FIG. 7 illustrates an overall block diagram of the Flash ADC;

FIG. 8 illustrates a prior art Flash ADC;

FIG. 9 illustrates a more detailed diagram of the comparator portion ofthe Flash ADC of the present disclosure;

FIGS. 10 and 10 a illustrate a block diagram of a comparator string;

FIG. 11 illustrates a timing diagram for the operation of the compareoperation;

FIG. 12 illustrates a schematic diagram of the bias circuitry for theresistor ladder;

FIG. 13 illustrates a schematic diagram for the first comparatorsection;

FIG. 14 illustrates a schematic diagram for the second comparatorsection;

FIG. 15 illustrates a schematic diagram for the reconfigurable latch;

FIG. 16 illustrates a gain response curve for the reconfigurable latch;

FIG. 17 illustrates a schematic diagram for the dynamic latch;

FIG. 18 illustrates a simplified block diagram of the PID;

FIG. 19 illustrates a more detailed block diagram of the PID;

FIGS. 20 a and 2 b illustrate a z-domain plot of amplitude and phase;

FIG. 21 illustrates a frequency plot of a low pass filter;

FIG. 22 illustrates a frequency response of the sinc filter;

FIGS. 23 a and 23 b illustrate a block diagram of one implementation ofthe PID;

FIG. 24 illustrates a Bode plot of the overall digital compensatorcomprised of the PID and LPF;

FIG. 25 illustrates a more detailed waveform of the sinc filter;

FIG. 26 illustrates a plot of the voltage response in a prior art systemto positive and negative transients;

FIGS. 27 a and 27 b illustrate voltage plots for transients in thepresence of voltage positioning for both low and high current,respectively;

FIGS. 28 a and 28 b illustrate the relationship between the voltage setpoint and the current level;

FIG. 29 illustrates a flow chart depicting the operation of voltagepositioning;

FIG. 30 illustrates a block diagram of the voltage positioning in thecurrent sensing operation utilizing two current sensors;

FIG. 31 illustrates a schematic diagram of the circuitry for determiningthe inductor current;

FIG. 32 illustrates a schematic diagram of the method for determiningthe capacitor current;

FIG. 33 illustrates a diagrammatic view of the method for measuring thetotal load current without Hall sensors;

FIG. 34 illustrates a diagrammatic view of the DPWM;

FIG. 35 illustrates a more detailed diagrammatic view of the DPWM;

FIGS. 36 a and 36 b illustrate a block diagram of the trim and limitsub-system;

FIG. 37 illustrates a block diagram of the DPWM timing register programmodel;

FIG. 38 illustrates a block diagram of the shut-down sources;

FIG. 39 illustrates a timing diagram for the sync operation;

FIG. 40 illustrates a timing diagram for the frame skipping operation;

FIG. 41 illustrates a simplified block diagram of the bypass logic;

FIG. 42 illustrates a flow chart for the operation of the patterngenerator for creation of the edges of the various phases;

FIG. 43 illustrates a flow chart for the operation of the u(n)selection;

FIG. 44 a is a functional block diagram of over current protectioncircuitry;

FIG. 44 b illustrates an integrator hold circuit responsive to theprimary interrupt;

FIG. 44 c is a flow diagram illustrating the operation of the integratorhold circuit of FIG. 44 b;

FIG. 45 is a timing diagram illustrating the operation of a phase outputof the digital pulse width modulator responsive to an over currentdetection signal;

FIG. 46 is a timing diagram illustrating the use of a blanking pulse;

FIG. 47 is a flow diagram illustrating the generation of primary andsecondary interrupts by the over current protection circuitry;

FIG. 48 is a flow diagram illustrating the operation of the resetcircuitry of the over current protection circuitry;

FIG. 49 is a functional block diagram illustrating the circuitry forproviding over voltage and over temperature protections for a digitalpulse with modulator;

FIG. 50 is a flow diagram illustrating the method for providing overvoltage and over temperature protections;

FIG. 51 illustrates a diagrammatic view of the PLL;

FIG. 52 a is a voltage diagram of an input voltage for a switched powersupply with an analog controller;

FIG. 52 b is a voltage diagram of an input voltage for a switched powersupply with the digital controller of the present invention;

FIG. 53 is a flow diagram of a method for controlling the input voltageduring soft start;

FIG. 54 is a functional block diagram of a switched power supply havingnonlinear operation;

FIG. 55 is a diagram illustrating the nonlinear operation of theswitched power supply of

FIG. 54;

FIG. 56 illustrates an improved switched power supply having linearoperation over all voltage ranges;

FIG. 57 is a diagram illustrating the linear operation of the switchedpower supply of FIG. 56; and

FIG. 58 is a flow diagram describing the operation of the linearlyoperating switched power supply of FIG. 56.

DETAILED DESCRIPTION OF THE INVENTION

Referring now to FIG. 1, there is illustrated a top level schematicdiagram for the switching power supply of the present embodiment, whichin this Fig. is illustrated as a half bridge power supply. The mainportion of the power supply comprises a primary switch group 102 that isoperable to receive an input voltage on a node 104, this being a DCvoltage, and ground on a node 106. The primary switch group 102 iscoupled through an isolation transformer 108 to a secondary switch group110. The secondary switch group 110 is operable to drive an outputvoltage node 112 that is connected to one terminal of a load 114, thesecondary switch group 110 also having a ground connection on a node116, the load 114 disposed between the node 112 and the node 116. Thetwo switch groups 102 and 110 are operable to operate in conjunctionwith various pulse inputs on a control bus 118 associated with theprimary switch group 102 and with various pulse inputs on a control bus126 associated with the secondary switch group 110.

A digital control circuit 124 is provided which is operable to controlthe operation of the primary switch group 102 and the secondary switchgroup 110. The nodes 104 and 106 are provided as inputs to the digitalcontrol circuit 124 for sensing the voltage and current on the primary,the digital control circuit 124 generating the information on the bus118 for control of the primary switch group 102. The control circuit 124must be isolated from the secondary switch group 110. This isfacilitated by driving a bus 126 through an isolation circuit 128, suchas an opto-isolator, to drive the bus 120. Similarly, the controlcircuit 124 is operable to sense the voltage and current levels on theoutput node 112 through sense lines 130 which are also connected throughan isolation circuit 132 to the digital control circuit 124. The digitalcontrol circuit 124 is also interfaced to a bus 136 to receive externalcontrol/configuration information. This can be facilitated with a serialdata bus such as an SMB serial data bus.

Referring now to FIG. 2, there is illustrated a detailed schematicdiagram of the primary switch group 102, isolation transformer 108 andsecondary switch group 110. The node 104 is connected to one side of thesource-drain path of a power switching transistor 202, the other sidethereof connected to a node 204. Node 204 is connected to one side ofthe primary of isolation transformer 108, a primary 206. The other sideof primary 206 is connected to a node 208. Node 208 is coupled to node104 through a capacitor 210. Node 106 is coupled to one side of thesource-drain path of a switching transistor 212, the other side thereofconnected to node 204. Node 208 is coupled through a capacitor 214 tonode 106. A diode 218 has the anode thereof connected to node 208 andthe cathode thereof connected to a node 220, node 220 connected to oneside of the source-drain path of a switching transistor 222, the otherside thereof connected to node 204.

Switching transistor 212 is controlled by a switching pulse P1, the gateof switching transistor 202 controlled by a switching pulse P2 and thegate of switching transistor 222 controlled by switching pulse P3.Switching pulses P1, P2 and P3 all form part of the bus 118.

The secondary switch group 110 is comprised of a switching transistor230 having the source-drain path thereof connected between the node 116and a node 232, the gate thereof controlled by a switching pulse P5.Node 232 is connected to one side of a winding 234 which forms part ofthe secondary of the isolation transformer 108. The other side ofwinding 234 is connected to a center tap node 236, node 236 connected toone side of a winding 238, the other side thereof connected to a node240. Winding 238 and winding 234 form the secondary of transformer 108.

Node 240 is connected to one side of the source-drain path of aswitching transistor 242, the other side thereof connected to node 116and the gate thereof connected to a switching pulse P4. An inductor 244is connected between node 236 and the output node 112. The output node112 is coupled to the ground node 116 through a capacitor 246 which isconnected proximate to the other side of the source-drain path oftransistor 230 and coupled through a capacitor 248 to node 116 proximateto the other side of the source-drain path of switching transistor 242.

Referring now to FIG. 3, there is illustrated a timing diagram forgenerating the switching pulses to operate the switch of FIG. 2. Theswitching pulse P1 is a pulse-width modulated switching pulse having arising edge 320. The rising edge 320 changes the level to a high level322 which then returns to the low level at a falling edge 324. Theswitching pulse P2 is delayed from the falling edge 324 by a delayt_(d1). The rising edge 326 changes the level of switching pulse P2 to ahigh level 328 followed by a change back to a low level having a fallingedge 330. The switching pulse P3 goes from a low level to a high levelahead of the falling edge of P2 by delay time t_(d2). The switchingpulse P3 returns to the low level at a falling edge 336.

In the output switch, the switching pulse P4 goes from a low level to ahigh level 336 at a rising edge 338. The rising edge 338 is delayed fromthe rising edge 320 by a delay t_(d3). The switching pulse P4 returns toa low level ahead of the falling edge of P1 by delay time t_(d3). Theswitching pulse P5 goes from a low level to a high level 342 at a risingedge 344 which is delayed from edge 326 of switching pulse P2 by a delayt_(d3). Switching pulse P5 returns to a low level ahead of the risingedge of P3 by delay t_(d3).

It can be seen that the switches 202 and 212 in FIG. 2 are controlled byswitching pulses P1 and P2. The delay t_(d1) is the duration of timerequired for transistor 212 to go from a conducting state to anon-conducting state and prior to transistor 202 going to a conductingstate. The delay t_(d1) is a delay that is required in order to ensurethat the switches are completely off such that connecting the node 204to the ground node 106 does not cause current to flow through transistor202. This could result in a “shoot-through” current spike. Dependingupon the circuit components and operating frequency, it may be necessaryto vary this delay. Similarly, transistor 222 will be turned on prior toturning off switch 202 with the delay t_(d2) allowing the diode 218 tobe placed in parallel with the primary 206 prior to turning offtransistor 202. Similarly, on the output switch, it is necessary thattransistor 242 is maintained in a non-conducting state until transistor212 is fully turned on and node 204 is sufficiently grounded. Further,it is necessary that the falling edge 346 be delayed until thetransistor 222 has fully turned on, which requires the delay t_(d3).This timing is conventional and, depending upon the application, thevarious delays will be adjusted, these adjustments due to the size ofthe load, circuit characteristics and operating frequency.

Digital Controller—Overall

Referring now to FIG. 4, there is illustrated a block diagram of thedigital controller 124 of FIG. 1. As described hereinabove, theswitching converter is generally realized with a half bridge converter,but a simpler buck converter 402 is illustrated in this figure. Thisrequires a plurality of phases 404 for controlling the switches internalto the buck converter 402. This will allow a DC input voltage to beconverted to a DC output voltage on output 406. The digital controllersenses the output voltage on the output 406 as a sense voltage,V_(SENSE), and inputs this to one input of a differential analog todigital converter (ADC) 408. The other input of the ADC 408 is connectedto an analog or reference voltage generated by a V_(REF) generator 410that, as will be described hereinbelow, comprises a digital to analogconverter (DAC).

The output of the ADC 408 is a digital output that represents thedifference between the analog output voltage on the DC output 406 andthe “set point” generated by V_(REF) generator 410. The output of theV_(REF) generator 410 is typically the desired output voltage. As such,the operation of the control loop at regulation will typically result ina “0” output from the ADC 408. As will be described hereinbelow, this isthe “0” code for the ADC 408. This is input to a digital compensator412, which is operable to provide some phase lead in the loop. The buckconverter 402 is comprised of a combination of a series inductor andshunt capacitor that forms an LC network, which provides a phase lag of180°. The control loop will typically be provided by a negative feedbackloop and will result in an additional negative phase shift of 180°. Ifthe loop were allowed to operate in this manner, this would result in a0° total phase change which would be an unstable loop. As such, thedigital compensator 412 provides some phase lead to stabilize the loop.The output of digital compensator 412 provides the digital control valueu(n) on a digital output bus 414 for input to a digital pulse widthmodulator (DPWM) 416. This provides the various clock signals whichprovide the switching phases 404 to the buck converter 402 (or to a halfbridge converter described herein above).

The ADC 408, digital compensator 412 and DPWM 416 are realized inhardware such that they provide relatively fast digital response and,once operating, operate in a fixed manner. However, each of the ADC 408,digital compensator 412, DPWM 416 and V_(REF) generator 410 are operableto be configured and have the operation thereof monitored. The V_(REF)generator 410 has a configuration block 420 associated therewith forconfiguring the operation thereof such that the voltage of the V_(REF)generator 410 can be controlled. Additionally, a monitoring circuit 422is provided for monitoring the operation thereof. Similarly, the ADC 408has a configuration block 424 for configuring the operation thereof anda monitoring block 426 for monitoring the operation thereof. The digitalcompensator 412 has a configuration block 428 for configuring theoperation thereof and a monitoring block 430 for monitoring theoperation thereof. The DPWM 416 has a configuration block 432 forconfiguring the operation thereof and a monitoring block 436 formonitoring the operation thereof.

As will be described hereinbelow, the ADC 408 is a parallel dataconverter that is configured with a Flash ADC topology. The digitalcompensator 412 is configured with a proportional-integral-derivative(PID) compensator with post processing filtering and DPWM 416 isrealized with a state machine. The PID compensator is a discretecompensation network that is operable to apply a discrete time PIDcontrol law to the signal. The operation of each of these blocks iscontrolled through the associated configuration and monitoring blockswith a microcontroller 440. The microcontroller 440 is an instructionbased engine that operates on instructions that can be downloaded toFlash memory 442, which is non-volatile memory. A serial data input 442allows instructions to be input to the microcontroller 440 for storagein the memory 442 and for various debug and control operations.Additionally, error handling is provided by a block 446 that basicallyprovides for over current protection and over voltage protection toprevent damage to the buck converter 402 under certain conditions, aswill be described in more detail hereinbelow.

By providing a digital controller that, when operating and configured,operates independent of the programmable microcontroller 440, thefunctionality of the digital controller is embedded primarily within thecircuitry of the primary block involving the ADC block 408, the digitalcompensator block 412 and the DPWM block 416. The microcontroller 440basically is the “housekeeper” for the digital controller which isoperable to monitor the operation thereof. When the digital controlleris operating at voltage regulation and once configured, very few actionsneed to be taken by the microcontroller 440. However, when the digitalcontroller is originally configured, depending upon the environment, thetype of switching converter utilized, etc., the digital controller willbe configured by the microcontroller 440 for a specific application.Even for the given application, there are certain transients that occur,such as when the converter is powered up, when short circuits occur,when transient loads are applied, etc. and, thus, certain parameters ofthe various blocks need to be varied to accommodate such during theoperation of the DC-DC converter. By providing an instruction basedengine such as the microcontroller 440 in a monitoring mode andconfiguration mode, the operation of the digital controller can bemonitored and then the parameters thereof changed temporarily, ifnecessary, to account for this change. To implement the entire digitalcontroller in an instruction-based engine such as a DSP would require alarge amount of programming operations. By providing a hardware baseddigital controller as the primary block, the functionality has beenembedded within the hardware by the chip designer. The DSP solution, onthe other hand, typically utilizes a general purpose DSP and the valueor functionality of the digital controller is facilitated throughprogramming, which can be complex and typically is utilized only forvery high-end digital controllers. Further, the implementation of theprimary digital control in hardware provides for a more efficient designthat utilizes the circuitry and is more power efficient, which isimportant in low power DC-DC converters, without sacrificing thebenefits of digital control.

Referring now to FIG. 5, there is illustrated a more detailed blockdiagram of the digital controller. The ADC 408 is a differential FlashADC that is operable to determine as a digital value the differencebetween the voltage on the DC output node 406, that being the V_(SENSE)voltage, and a reference voltage on a node 502. This analog referencevoltage on node 502 is generated by the V_(REF) generator 410. This iscomprised of an analog reference voltage generator 504 which is operableto generate a fixed analog reference voltage based on an internalreference such as a bandgap generator. The bandgap generator is aconventional circuit that is utilized to generate temperature andprocess stable voltages. This is not shown in the illustration of FIG.5. The V_(REF) generator 504 will generate this reference voltage andprovide it as a reference input to a conventional reference digital toanalog converter 506 (reference DAC). This is a scaling DAC that isoperable to receive a digital word on a bus 508 from a reference DACcontrol block 510 that is controlled by the microcontroller 440. This isbasically a register that can be written to for the purpose ofgenerating the reference DAC voltage. The reference DAC 506 is operableto convert this digital value on bus 508 to an analog voltage on node502 for input to one of the differential inputs of the ADC 408.Typically, the voltage generated by V_(REF) generator 504 is a 1.25 Vanalog voltage. The output of the reference DAC 506 comprises thedesired voltage of the DC-DC converter. In one embodiment, this isapproximately 1.0 V, a conventional processor voltage. The referencevoltage on node 502 is compared with the V_(SENSE) voltage on node 406and, when regulated, this should essentially be zero. In the test modeof operation, there is provided a switch 512 which is operable to shortthe two inputs together. This will be described hereinbelow.

The ADC 408, as will be described hereinbelow, is a parallel ADC of theFlash type. It is a window ADC that is operable to generate a zerovoltage output when the differential input is “0.” An ADC control block514 is operable to provide a control input to the ADC 408. The controlblock 514 provides a variable LSB input to the ADC 408 for use with someof various features thereof. The ADC operates on an ADC CK clock signaland also generates an end of conversion cycle interrupt, EOC1 IRQ. Thisprovides an indication of when a data conversion operation is completeon a given sample and digital data associated with the analog sample isready to be output. The data is output through an inverter circuit 516for input to one input of a 4-input digital multiplexer 518, which ispart of the input interface to the digital compensator 412.

The digital compensator 412, in addition to receiving the output of theADC 408 through the inverter 516, is also operable to receive a groundinput on a digital input bus 520, ADC data from a register 522 through abus 524 for digitally generated ADC data, primarily for test purposes,and also a “raw” data input on a bus 526. In one mode of operation,primarily associated with start-up and the such, the sensed voltage,V_(SENSE), is determined by another ADC, which is described hereinbelow,which is a SAR ADC. This is a slower ADC and the output thereof isstored in a special function register, V_(SENSE/SFR), the output ofwhich is provided on a bus 528. The difference between the digitalrepresentation of the V_(SENSE) voltage and the actual input toreference DAC 506 on the bus 508 is determined by a digital subtractionblock 530, the output of which comprises the bus 526. Therefore, asingle-ended SAR can be utilized to bypass the ADC 408 and determine avalue for input to the digital compensator 412 during start-up and thesuch, this providing the differential operation in the digital domain.However, during regulation, the ADC 408 is the preferred input dataconverter.

One situation in which the difference between the digital representationof the output voltage V_(SENSE) and the control parameters within thereference DAC 506 is used for controlling the voltage output is during asoft start procedure. During the soft start procedure the input voltageis slowly ramped from zero up to the set point of the power supply. Whenthe digital controller is in normal operating mode, the voltage errorsignal generated by the ADC 408 is provided from the output ofmultiplexer 518 to the input of the PID 540. The ADC 408, as describedherein above, generates the voltage error signal by making a comparisonbetween the output voltage signal V_(SENSE) and the reference voltagesignal applied to the analog to digital converter 408. This generationof the voltage error signal occurs in the analog domain.

During a soft start process, it is desired to more particularly controlthe manner in which the voltage error signal ramps up from zero to theset point voltage. This is achieved by creating a voltage error signalon bus 526 which is applied by multiplexer 518 to the input of the PID540. The voltage error signal generated on the bus 526 is generatedcompletely in the digital domain using the digital subtraction block 530and is under control of the microcontroller 440. A first input isprovided to the negative input of the digital subtraction block 530 onbus 528 from the V_(SENSE)/SFR 670 which will be more fully describedwith respect to FIGS. 6 a-6 b. The V_(SENSE)/SFR register 670 stores adigital version of the V_(SENSE) signal that is generated by the 12-bitSAR ADC 660 within the microcontroller 440. This analog to digitalconverter 660 is used because it has greater common mode range than thatof the analog to digital converter 408 outside of the microcontroller440. The voltage error signal provided on bus 526 is controlled by acontrol parameter provided from the reference DAC control register 510.The control parameter written to the reference DAC control 510 isgenerated and controlled by the microcontroller 440. In this manner, thevalue of the voltage error signal generated on bus 526 may be controlledby controlling the control parameters stored in the reference DACcontrol 510 by the microcontroller 440.

By controlling the generation of the voltage error signal applied to thedigital controller, the input voltage of the switched power supply maybe controlled to any level from zero to its set point. Analogcontrollers may only control the slope of the input voltage during softstart in the manner illustrated in FIG. 52 a. In this figure, the inputvoltage is controlled to proceed from a zero value 5202 up to a setpoint value at 5204 along a continuous ramp. The only thing that can becontrolled when proceeding from point 5202 to point 5204 is the slope ofthe ramp. By changing the value of an external capacitor, the slopebetween point 5202 and 5204 may be controlled in an analog controller.

Referring now however to FIG. 52B, within a digital controller accordingto the present invention, the microcontroller 440 may establish variouscontrol parameters within the reference DAC control register 510 suchthat the voltage error signal applied on bus 526 causes the inputvoltage of the switched power supply to proceed from the zero value at5206 to the set point value 5208 in a plurality of voltage input waveform profiles. In this example, the input voltage wave form proceeds toa point 5210 and remains at this level until point 5212. The inputvoltage next increases to a second voltage level at point 5214 andremains at the second level until it increases onward to the set pointvoltage at 5216. The illustration in FIG. 52B illustrates only a singleinput voltage profile achievable using the above-described system andmethod and it should be readily apparent that any number of inputvoltage profiles may be achieved by the microcontroller 440 controllingthe value stored within the reference DAC control register 510.

FIG. 53 illustrates one method in which the microcontroller 440 couldcontrol the voltage error signal provided to multiplexer 518 during asoft start procedure. At inquiry block 5302, the microcontroller 440monitors for the occurrence of a soft start condition. If no soft startcondition is detected, inquiry step 5302 continues to monitor for thesoft start condition. Once a soft start condition is detected, themicrocontroller 440 switches, at step 5304, the multiplexer 518 toprovide the digital voltage error signal from the reference DAC and theV_(Sense)/SFR register of the microcontroller. The microcontroller 440initially stores, at step 5306, a control parameter within the referenceDAC 510 to initiate the soft start process. A voltage error signal isgenerated at step 5308 from this initial control parameter. Inquiry step5310 determines whether the soft start process has been completed. Ifthe soft start process has not been completed, a next control parametermay be stored by the microcontroller at step 5312. As discussedpreviously, once a particular input voltage level is achieved during thesoft start process, the input voltage may be maintained at this level byproviding the appropriate voltage error signal for as long as desired.Control passes back to step 5308 wherein the appropriate voltage errorsignal is generated for the newly provided control parameter at step5312. Once inquiry step 5310 determines that the soft start process hasbeen completed, the microcontroller 440 switches the multiplexer 518such that the analog voltage error signal is applied from the ADC 408.Using the above-described method, the microcontroller 440 is able tocontrol the voltage profile for the input voltage of the power supply bycontrolling the error signal applied to the controller.

Referring now to FIG. 54, there is illustrated a functional blockdiagram of the fast feedback loop between the output of the switchedpower supply 5402 and the flash ADC 408. The flash ADC 408 compares theoutput voltage V_(Sense) with a reference voltage provided from thedigital to analog converter 506 that converts an analog signal V_(REF)into an analog representation. The comparison of the output voltageV_(Sense) with the reference voltage V_(REF) provides a voltage errorsignal which is provided to the digital compensator 412. The digitalcompensator 412 provides the digital control value u(n) to a digitalpulse width modulator 416. The digital pulse width modulator 416, asdescribed previously, provides the clock signals which provide theswitching signals to the switched power supply 5402. The output of theswitched power supply 5402 is connected to the ADC 408 to provide thevoltage error signal by comparison of the output voltage V_(Sense) tothe reference voltage V_(REF).

The ADC 408 provides a relatively fast digital response. A fast ADC suchas the ADC 408 described hereinabove has common mode limitations thatlimit its ability to convert signals of lower voltages and stilleffectively provide a linear response within these lower voltageregions. However, a lower voltage reference is often needed forcontrolling functions such as soft start and power factor correction.For the system in FIG. 54, during closed loop operation, the outputvoltage signal is fed into the fast differential ADC 408 wherein it iscompared with the reference voltage V_(REF) provided from the DAC 506.The resulting error is provided to the digital compensator 412. As canbe seen in FIG. 55, the digital analog converter 506 may only beadjusted within the common mode range 5502. The region 5504 outside ofthe common mode range 5502 provides a nonlinear response. Sweeping theDAC 506 in a stable loop would yield the response illustrated in FIG. 55consisting of the common mode range 5502 which provides a linearresponse and the nonlinear region 5504 occurring at the lower voltageranges.

Referring now to FIG. 56, there is illustrated a functional blockdiagram of a nonlinear switching power supply capable of performing alinear operations in the lower voltage range. A fast feedback loop isprovided between the output of the switched power supply 5402 and one ofthe inputs of fast analog to digital converter 408. The analog todigital converter 408 compares the V_(Sense) signal from the output ofthe switched power supply 5402 and compares this with an analogrepresentation of a corrected voltage reference signal provided by thedigital to analog converter 506. The ADC 408 provides a voltage errorsignal to the digital compensator 412 which generates the control signalu(n) that is provided to the digital pulse width modulator 416. Thedigital pulse width modulator 416 provides switching signals to driveswitches within the switched power supply 5402.

The solution to linearizing the nonlinear portion 5504 of FIG. 55 is tointroduce a slow feedback loop into the system that injects an errorsignal into the reference voltage V_(REF) applied to the digital analogconverter 506. The slow feedback loop is achieved by providing theoutput voltage V_(Sense) to the input of the 12-bit SAR analog todigital converter 660 within the microcontroller 440. The output of theADC 660 is filtered via a digital low pass filter 5602 implementedwithin software of the microcontroller 440. The filtered signal has itsderivative taken at block 5604. The derivation is also implementedwithin software of the microcontroller 440. The derivative is applied toan adder circuit 5608 such that the derivative is injected into thereference voltage signal V_(REF) at the adder 5608 as an error offset.This corrected voltage reference signal is applied to the input of thedigital to analog controller 5604. The digital to analog converter 5604converts the corrected voltage reference signal to analog format forapplication to the ADC 408.

This second slow control loop has a lower bandwidth than the fastcontrol loop and effectively regulates the slope of the voltage outputfrom the switched power supply 5402. The slow ADC 660 has a much higherresolution and is monotonic over its operating range. However, the slowADC 660 has a relatively slow response. The slower control loop adds asufficient error to the V_(REF) signal applied to the digital to analogconverter 506 to overcome the nonlinearity in the signal provided by thefast ADC 408 at low voltage ranges. The monotonicity of the slower ADC660 ensures good linear control over the lower voltage range.

Thus, using the circuit illustrated in FIG. 56, a voltage versus timeresponse such as that illustrated in FIG. 57 may be achieved. As can beseen, the voltage response over all voltage ranges is linear over theentire voltage range rather than only being linear within the commonmode range and nonlinear within the lower voltage range as wasillustrated in FIG. 55.

Referring now to FIG. 58, there is illustrated a flow diagram describingthe operation of the slow feedback control loop. The process begins bydetecting the output voltage from the switched power supply 5402 at step5802. The slow control loop uses the analog to digital converter 660 toconvert the analog output voltage signal into a digital signal at step5804. The digital representation of the output voltage signal is lowpass filtered at step 5806 by the digital low pass filter 5602. Thefiltered signal has its derivative taken at step 5808. The determinedderivative is injected into the voltage reference signal V_(REF) as anoffset at step 5810. This corrected V_(REF) is used to generate ananalog representation of the corrected V_(REF) signal at step 5812 fromthe digital to analog controller 5812. The corrected voltage referencesignal is applied to the analog to digital converter 408 and comparedwith the output voltage V_(Sense) to generate a voltage error signal atstep 5814 for application to the digital compensator 412.

The output of the multiplexer 518 is input to a PID controller block,which provides a proportional, integral, derivative (PID) controlalgorithm. One difficulty associated with designing a controller arisesfrom the LC resonance of a buck converter. An open-loopfrequency-response analysis exhibits a resonant peak at the cutofffrequency of the LC filter. A sharp peak, quantified by the qualityfactor (Q), is desirable for efficient power conversion for losslesspower conversion. For a simple integral control, this resonant peak mustbe kept below unity gain in the open-loop frequency response to ensurestability. Such a controller configuration has a low loop bandwidth andleads to slow transit response characteristic. This PID block 540provides the requisite loop stability without sacrificing bandwidth andimproves the loop's transient response. The proportional and derivativecontrol blocks, as will be described hereinbelow, introduce compensationzeros that push unity-gain beyond the resonant peak and eliminates thebandwidth limitation otherwise imposed by the resonant nature of thebuck converter. There is provided a PID control block 542 that controlsthe operation of the PID 540 by providing, as will be set forthhereinbelow, gain constants for the operation thereof. The operation isclocked with a filter clock, FILTCLK, on a clock input 544. The input tothe PID 540 is determined by the output of multiplexer 518, which iscontrolled by a PID input control block 546. The clock rate is around 10MHz, wherein the switching frequency of the power supply is around 500kHz

The analog corollary to the digital controller has one inherent benefitin that the overall operation of the analog controller has an inherentlow pass filter function associated therewith. The PID 540, on the otherhand, has an amplitude and phase response that increases with increasingfrequency such that the gain thereof becomes relatively high at higherfrequencies and the phase also increases in an ever increasing phaseleading manner. To accommodate the frequency response of the PID, postprocessing filtering is required. This is facilitated in the presentembodiment with either a low pass filter, represented by an LPF filterblock 550 or a sinc filter block 552. The output of the PID 540 is inputto both of these blocks 550 and 552 and the outputs thereof selectedwith a two-input digital multiplexer 554. The sinc filter operation 552provides for a plurality of “notches” which are controlled by a sinccontrol block 556, the sinc filter block 552 clocked by the FILTCLKclock signal. The LPF filter block 550 also utilizes variable poles andzeros that are set by an LPF control block 558. The LPF filter block 550is also clocked by the filter clock, FILTCLK. The output of multiplexer554 provides the output from the digital compensator 412, the outputselected by the multiplexer 554 controlled by a filter select block 560.

The output of the multiplexer 554 from the digital compensator 412 isprovided on a digital data bus 562. This is input to a PID data register564 for the purpose of monitoring the operation thereof, such that theoutput of the digital compensator block 412 can be monitored. The outputof the multiplexer 554 is also input to the input of a two-input digitalmultiplexer 566, the other input thereof receiving data from the PIDdata block 564, such that the operation of the compensator 412 can bebypassed. The multiplexer 566 is controlled by a DPWM input controlblock 568. The output of the multiplexer 566 provides the u(n) errorsignal, which is output on a bus 570 to the DPWM 416. The DPWM 416, asset forth hereinabove, is a state machine and is controlled by a DPWMcontrol block 572. The DPWM block, as will be described hereinbelow, isoperable to receive various control signals from the DPWM control block572 from the microcontroller 442 and is also operable to generate aplurality of interrupts (not shown) and receive various interrupts. Forexample, at the end of a given frame, there will be an EOFIRQ interruptgenerated, and the DPWM 416 will also receive various interrupts fromthe error handling block 446 to indicate either over current situationsor over voltage situations.

Referring now to FIGS. 6 a-6 b, there is illustrated a detailed blockdiagram of the microcontroller 440. This microcontroller 440 is an 8051instruction-based engine which is substantially disclosed in U.S. patentapplication Ser. No. 10/244,344, filed on Sep. 16, 2002 and entitled“Precision Oscillator for an Asynchronous Transmission System,” which isincorporated herein in its entirety by reference for all purposeswhatsoever. At the center of the microcontroller 440 is a processingcore 602 which is an 8051 microprocessor engine. This is aninstruction-based engine. There is provided a 32K byte Flash memoryblock 604, 256 byte IRAM block 606 and a 1K byte XRAM block 608,providing memory for the processing core 602. Clock signals are providedto the core 602 in the form of a system clock, SYSCLK, on a clock line610. This is provided on the output of a multiplexer 612. Themultiplexer is operable to receive the input thereof from a 20 MHz bootoscillator block 614, an input from an 80 kHz low frequency oscillatorblock 616 to provide an 80 kHz clock for use in a sleep mode, or ahigher frequency clock in the form of a divided down 25 MHz oscillator618. The 25 MHz oscillator is the primary oscillator at the operatingfrequency of the core 602, as the core 602 operates at high frequency orat low frequency. However, at low frequency, the processing ofinstructions occurs at a much slower rate and this mode is typicallyused in a sleep mode. In the normal operating mode, typically the higherfrequency clock oscillator is utilized. This clock is a non-crystalbased clock and has an accuracy of approximately 2%. The output of theclock 618 is input through a two-input multiplexer 620 to themultiplexer 612, the output of multiplexer 620 passed through a divideblock 622 in order to divide the frequency of the clock, if necessary.Additionally, an external clock is input to the other input ofmultiplexer 620, such that either the internally generated 25 MHz clockcan be utilized or an external clock can be utilized. A phase lock loop624 is provided which is controlled by a PLL control block 626 and thisutilizes the 25 MHz clock 618 as a reference and then multiplies thisclock up to as high as 400 kHz. This provides an output to one end ofthe multiplexer 612 for selection as the SYSCLK. This PLL 624 isoperable to generate the other clocks associated with the operation of adigital controller, the clock for the DPWM 416, PWMCK, the filter clock,FELTCLK, and the ADC clock, ADCCLK. This will be described hereinbelow.

The core 602 is also operable to receive a Reset signal on a block 630,which is operable to generate a reset when it is not in a debugoperating mode. In a debug operating mode, the Reset input on a node 631is input to the clock input of a debug hardware block 634 to provide aclock signal thereto, the other input being a serial data input on aline 635. This is a two-wire serial data port that allows for very lowclocked data to be input to the core 602 during a debug mode. In thereset mode, the reset block 630 provides the reset signal to the core602.

The core 602 is interfaced through a special function register (SFR) bus630 to various I/O blocks. In the embodiment illustrated herein, fourtimers 632 are provided. Each of these timers is operable to have theparameters thereof set, and initiated and each of them generates varioustimer interrupts, TMRXX IRQ, signals. Additionally, there are provided anumber of serial bus configurations for allowing for various formats ofa serial data interface. One of these is the SM Bus/I2C format, in ablock 634. This is a conventional serial data format. Additionally,there is provided a UART functionality in a block 636. There is provideda programmable counter/timer array (PCA) block 638 and a plurality ofport latches 640 for interfacing with a port “0” block 642 and a port“1” block 644 for transmitting and receiving data therefrom. All of theblocks 632-640 are interfaced through a crossbar matrix block 646, whichis disclosed in U.S. Pat. No. 6,738,858, issued May 18, 2004, which isincorporated herein by reference. The crossbar matrix is operable toselectively connect any of the outputs of the blocks 632-640 to any of aplurality of output pins associated with the port driver 642 and 644,there being eight pins 650 associated with the port “0” driver 642 andeight pins 652 associated with the port “1” driver. These pins canfunction as digital outputs, digital inputs or analog inputs.

For analog sensing, all of the eight pins 652 associated with the port“1” driver are connectable to analog inputs of a multiple input analogmultiplexer 656 which is operable to receive eight analog inputs, AIN0,AIN1, . . . , AIN7, a V_(SENSE) input and a Temperature input. The inputvoltage is connected to the AIN0 input for sensing thereof. A separatededicated pin is provided for the V_(SENSE) input for input to themultiplexer 656. An additional input is provided by an internaltemperature sensor 658, which senses the chip temperature, whichbasically constitutes the environmental temperature, this being an inputto the analog multiplexer 656. The output of the analog multiplexer 656is input to the input of a 12-bit SAR ADC 660, operating at a samplingclock of 500 Ksps. This is a single-ended ADC that provides the digitaloutput on a bus 662. The control for the ADC 660 is provided by the ADCcontrol block 664. The analog multiplexer 656 is controlled by an autoscan block 666, which is operable to scan through all of the inputs in acyclical manner. At the end of each conversion cycle, there is generatedan interrupt EOC0 IRQ indicating the end of the conversion cycle for theADC 660. This is input to the auto scan block 666 which will thenincrement the select control on the multiplexer to the next input toinitiate a second or subsequent conversion operation. For each scanstep, the output of the ADC 660 is “steered” or directed toward anassociated special function register (SFR)/limiter (LIM). Each of theseSFR/LIM blocks is operable to store the associated output, compare itwith an internal fixed upper and/or lower limit, which can be variedupon power-up, and then output an interrupt if it exceeds the limit(s).In the first five SFR/LIMs, there is provided an ADC window interrupt inan SFR/LIM block 668, an SFR/LIM block for the V_(SENSE) output 670, anSFR/LIM block 672 for the AIN0 output, an SFR/LIM block 674 for the AIN1input, and an SFR/LIM block 676 for the AIN2 input. The SFR/LIM blockfor the V_(SENSE) output 670 is used as the input on bus 528 to thedigital subtracter block 530 to determine the input to the digitalcompensator 412 during a startup or soft start process. Each of theseblocks 668-676 provide an associated interrupt, ADC0WINTIRQ, V_(SENSE)IRQ, AIN0VIN IRQ, AIN1 IRQ, and AIN2 IRQ. Since the core 602 can onlyhandle a certain number of interrupts, the remaining inputs, AIN3-AIN7and TEMP are associated with respective SFR/LIM blocks 678. The outputof each block 678 provides an associated interrupt to an OR gate 681.The output of the OR gate 680 provides an interrupt, which whenrecognized by the core 602, requires that the core 602 then “poll” theoutputs of the SFR/LIM blocks 678, it being recognized that each of theSFR/LIM blocks occupies a unique address in the address space of thecore 602, such that the contents thereof can be read, or in certaincircumstances, written to. Whenever an interrupt is generated, the core602 initiates an interrupt sub-routine for servicing that particularinterrupt, as is the case with any interrupt generated.

There is also provided a comparator function for generating a comparatorinterrupt. A comparator block 680 is provided which is operable to haveone compare input interface with the even ones of the pin 652 and asecond input interface with the odd inputs thereto. This is a fourcomparator block, which is controlled by a comparator control block 682and will generate a comparator interrupt whenever any of the respectiveinputs exceeds the threshold set therein.

Referring now to FIG. 6 c, there is illustrated a diagrammatic view ofan integrated circuit 690, which is operable to provide all of thefunctions for the digital control operation in a single integratedcircuit. This integrated circuit 690 requires only connections fromV_(SENSE) on a pin 692, switching control signals on output pins 693, apower supply input on a power supply pin 694 and a ground connection ona pin 695. With these minimal number of pins, the entire digital controloperation can be facilitated. This assumes that a program is provided inthe memory 442. If the program is not “hard coded,” some type of serialconnection on at least one pin 696 is required, but it should beunderstood that other pins in the system can be multiplexed for use inprogramming, since programming is facilitated in a nonoperating mode.Further, there are provided a plurality of pins 697 that are operable toreceive other sense analog input voltages. However, for thestraightforward operation of the digital controller, all that isrequired is the V_(SENSE) input. The other inputs are required for suchthings as over voltage protection and over current protection and fordetecting the peak current for the purposes of voltage positioning, aswill be described hereinbelow.

As set forth hereinabove, the digital control section is a hardwaredigital control section comprised of the ADC 408, the digitalcompensation network 412 and the DPWM 416. Once these blocks areparameterized, they will provide the control function associatedtherewith. The internal reference generator 410 is operable to providethe internal reference, for conversion to an analog signal by the DAC506. Thus, all the voltage reference information is contained in theintegrated circuit 690. The on chip self-contained microcontrollerprovides the monitoring and control functions such as over currentprotection, voltage positioning, etc. and, in general, provides allhousekeeping functions to monitor the operation of the hardware digitalcontrol stream. The self-contained clock and on-board memory provide forthe timing functions and the instructions for use by themicrocontroller, respectively. Therefore, it can be seen that the systemof the present disclosure provides for a single monolithic solution thatis low power due to the use of a state machine-driven digital controllerwithout requiring the power overhead of an instruction based system, butstill retains the intelligence of an instruction based system in themonitoring and reparameterizing aspect provided by the microcontroller440.

Flash ADC

Referring now to FIG. 7, there is illustrated a logic diagram of thewindow ADC 408. A first reference voltage is generated by an on-chipbandgap generator, a voltage V_(BG). The bandgap generator is aconventional circuit that combines a very stable voltage that is stableover temperature. This voltage is input to the voltage follower circuitcomprised of an amplifier 702, the output thereof driving the gate of ap-channel transistor 704. The source/drain path of the transistor 704 isconnected between V_(DD) and a node 708. Node 708 is connected to theother input of amplifier 702, such that the amplifier 702 and transistor704 provide a source follower configuration. Node 708 is connected to astring 710 of resistors of value “5R.” The output of amplifier 702 alsodrives a current mirror, such that the current through resistor string710 is mirrored over to the current mirror. The current mirror iscomprised of a p-channel transistor 712 and the gate thereof connectedto a node 714, node 714 connected to the output of amplifier 702. Thesource/drain of transistor 712 is connected between V_(DD) and a node728. Node 728 is connected to one side of the source/drain path of ann-channel transistor 716, the other side thereof connected to ground.The gate and drain transistor 716 are connected together to node 728 toform a diode-connected configuration. Node 714 is also connected to avariable width p-channel transistor 718, the source/drain path thereofconnected between V_(DD) and a node 720. Transistor 718, as will bedescribed herein below, is comprised of a plurality of parallelconnected binary-weighted transistors, the connection thereof beingprogrammable, such that one or all of the parallel connected transistorscan be connected in parallel on a selective basis.

Node 720 is connected on one side thereof to a resistor string comprisedof a plurality of resistors 722. There are provided sixty four of theseresistors 722 having a total resistive value of “R,” each having avoltage disposed there across equal to the voltage of a leastsignificant bit (LSB) of the ADC. This will be described in more detailherein below. The bottom of the resistor string of resistors 722 isconnected to a node 724, which is connected on one side thereof to thedrain of a variable n-channel transistor 726, the source thereofconnected to ground, and the gate thereof connected to the gate oftransistor 716 on a node 728. Transistor 726 is substantially identicalto transistor 718 and is also programmable to allow selection of thenumber of transistors connected together, which will be described inmore detail herein below.

A voltage input on an input node 730 represents the negative inputvoltage. This is input to one input of a unity gain amplifier 732, whichhas the other input thereof connected to the output on a node 734. Node734 represents the mid-point of the resistor string of resistors 722,such that there are an equal number of resistors above as below. Thus,for the disclosed embodiment of sixty four resistors 722, there will bethirty two resistors above and thirty two resistors below the point 734.The unity gain amplifier 732 provides the drive voltage node 734 andisolates the input voltage on node 730 therefrom.

The current through resistor string 710 is ratiometrically related tothe current through transistors 718 and 726 and all of the resistors722. Thus, the current through resistors 722 is set by the currentthrough resistor string 710, which current is set by the voltage on theinput to amplifier 702, voltage V_(BG), such that the current isV_(BG)/5R. The only way to vary the current of the resistors 722 isthrough the ratio of the size of the transistors 718 and 726 to the sizeof the transistor 704. This will be described in more detail hereinbelow.

Each of resistors 722, at the bottom thereof, is connected to one ofsixty four comparators on one input thereof of comparators 740, on oneinput thereof. (It is noted that the number sixty four defines a“window,” but any number of comparators could be utilized to representthe entire Flash ADC window). The other input of each of the comparators740 is connected to a node 742, which is connected to the positive inputvoltage V_(IN+). Therefore, the output of each of the respectivecomparators will be a “0” if the input voltage is below the resistor tapvoltage and a “1” if the input voltage is above the associated tapvoltage. The outputs of all of the comparators 740 having the referenceinput connected to resistor taps below the input voltage will have a “1”on the output thereof. This, therefore, represents a thermometer code onthe output thereof. This is input to a decoder 746 to decode thethermometer code and provide the digital output therefrom.

The output voltage from decoder 746, D_(OUT), represents the differencevoltage between the voltage on node 742 and the voltage on node 730,V_(IN+)-V_(IN−). By comparing the positive input voltage on node 742 tothe negative input voltage on node 730, the output voltage, V_(OUT),will have a resolution defined by the voltage across each of theresistors 722, this being the LSB of voltage. This overall circuitprovides the circuitry of the Flash ADC, this being a “window” Flash ADCas opposed to an absolute value ADC. When the difference between thevoltage on positive input voltage node 742 and negative input voltagenode 730 is “0,” the comparators 740 below the node 734 will have a “1”on the output thereof and the comparator 740 having the reference inputthereof connected to node 734 will have a “0” on the output thereof.This, as will be described herein below, represents the “0” code for theFlash ADC, this being a differential input ADC. As the size of thetransistors 718 and 726 is varied, this will vary the current throughthe resistors 722 and, therefore, vary the size of the LSB. However, the“0” code will not vary. In effect, the negative input voltage on node730 represents the reference voltage input of the ADC whereas thepositive input voltage on node 742 represents the analog input voltage.

To distinguish the current architecture of the Flash ADC with aconventional architecture, the prior art Flash ADC of FIG. 8 will bedescribed. In FIG. 8, a four comparator Flash ADC is described. Areference voltage is defined that is variable, this being for thepurpose of varying the size of the LSBs. This reference voltage isprovided on a node 802 at the top of a resistor ladder comprised of aplurality of tapped resistors 804. At each of the taps, there is anoutput provided to the reference input of an associated comparator 806.The other input on each of the comparators 806 is connected to an inputnode 808. For a single ended input, the reference voltage on node 802will typically be connected to the supply voltage and resistor 804adjusted such that the full rail-to-rail voltage could be provided. Inthis example, this would only provide a resolution of ¼ of the supplyvoltage. Typically, a very large number of comparators 806 will beprovided associated with a large number of resistors. For a 16-bit FlashADC, this would require 2¹⁶ comparators and a corresponding number ofresistors. This results in a significant power consumption for each ofthe comparators. However, for a differential input signal, it is onlynecessary to resolve the difference between a positive and negativeinput signal over a defined range. Thus, a smaller reference voltage canbe utilized which is divided by a predetermined number of resistors inthe corresponding comparator 806. In a prior art embodiment, thedifferential input voltage is determined by a differential amplifier 810receiving the positive and negative input voltage and outputting adifferential voltage on node 808. This differential voltage is theninput to the input of each of the comparators 806. Of course, in orderto utilize the full range, the output of the amplifier 810 must becentered around some common node voltage which is equal to V_(REF)/2. Inone alternate embodiment, the prior art system of FIG. 8 can have theLSB is changed by a factor of, for example, 10X, which will require thecommon mode voltage, V_(CM)=V_(ref)/2, to change by a factor of 10X.Although this will provide a stable zero code, the common mode voltage,V_(CM), of the amplifier 810 should be around V_(CM)/2 in order to havea large voltage swing.

It can be seen that, if the LSB size is varied through a variation ofthe reference voltage, this will cause the reference voltage on thezero-code node to change. If, for example, a node 820 associated withthe second from the top comparator 806 on the reference input thereofrepresents the zero-code wherein the positive input voltage equals thenegative input voltage, then, when the positive input voltage equals thenegative input voltage, this comparator will have a “0” on the outputthereof, comparators above will have a “0” output and comparators belowwill have a “1” output. As long as the voltage difference is “0,” andthe reference voltage is not varied, then the zero-code will not changebut, if the voltage V_(REF) is changed, the size of the LSB will changeand the zero code will also change, since the zero-code is now “coupled”to the value of V_(REF). Therefore, if the LSB is required to bechanged, then the tap associated with the resistor string that definesthe zero-code may change. This will be described in more detail hereinbelow.

Associated with each of the inputs of the comparator 806, is adistributed capacitance, which distributed capacitance would sum up to atotal capacitance of C_(T), represented by capacitor 814. It can be seenthat the amplifier 810 must drive the capacitance 814 during aconversion operation. By reducing the number of comparators in the“window,” the value of C_(T) can be reduced, in addition to the powerconsumption. However, the amplifier 810 must still drive this input witha capacitance.

Referring now to FIG. 9, there is illustrated a simplified diagram ofthe disclosed ADC of FIG. 7, which is utilized for comparison therewithto the prior art embodiment of FIG. 8. In this embodiment, it can beseen that the resistive string comprised of the resistors 722 are drivenby an upper current source 902 from the supply voltage, V_(DD), and thebottom of the resistive string is driven with a lower current source904. Both of these current sources provide a current I_(REF), which isvariable. This variable current source varies the current through theresistors 722 and, therefore, sets the size of the LSB or, morespecifically, the resolution of the ADC. The voltage on the node 734 isa negative input voltage and this provides the center reference voltageof the window with the current sources 902 and 904 in conjunction withthe current through the resistors, providing the LSB voltage incrementsincreasing toward current source 902 and decreasing toward currentsource 904. As the voltage on node 734 varies, the voltage across noderesistors 722 does not vary, as that voltage is controlled by thecurrent sources 902 and 904. However, if the current value of thecurrent sources 902 and 904 is varied, then the size of the LSB voltagewill vary.

As will be described herein below, each of the current sources 902 and904 are identical and are comprised of four separate parallel connectedcurrent sources, each having a binary-weighted current there through,such that a binary word can be input thereto for defining the valuethereof. In the disclosed embodiment, there are provided four currentsources, a 1x current source, a 2x current source, a 4x current sourceand an 8x current source, associated with a 4-bit word. This, however,is not meant to be limiting in that any number of current sources couldbe utilized, and any type of variable method for varying the currentsource could be utilized.

The output voltage, V_(OUT) is defined in the following equation:D _(OUT)=(V _(IN+) −V _(IN−))G

The value of G is related to the inverse of LSB as follows:

$G = \frac{1}{{LSB}\mspace{14mu}{size}}$The current through the resistor string is a ratiometric current suchthat it is the current through the resistor string 710 multiplied by aratio metric factor α. Thus, the current through the resistor string ofresistors 722 provided by transistors 718 and 726 is:

$\frac{V_{BG}}{5R}\alpha$where:

-   R is the total value of the sixty four resistors 722 in the ladder;    and-   α is a scaling or ratiometric factor.    Thus, the LSB is defined as the current through a given resistor and    it will be multiplied by the current through the resistor string    multiplied by the value of resistor, R, as follows:

${\left( {\frac{V_{BG}}{5R}\alpha} \right)\frac{R}{K}} = {\frac{V_{BG}}{5K}\alpha}$where:

-   K is a factor representing the number of resistors 722 in the    resistor string, there being sixty four in the disclosed embodiment.

As noted herein above, the ratio metric multiplier is a binary weightedmultiplier that, in the disclosed embodiment, utilizes a 4-bit word.This will be defined by the following relationship:

${LSB} = {\left( \frac{V_{BG}}{5K} \right) \cdot \left( \frac{{2^{3} \cdot {b3}} + {2^{2} \cdot {b2}} + {2^{1} \cdot {b1}} + {2^{0} \cdot {b0}}}{2} \right)}$${{where}:\alpha} = \left( \frac{{2^{3} \cdot {b3}} + {2^{2} \cdot {b2}} + {2^{1} \cdot {b1}} + {2^{0} \cdot {b0}}}{2} \right)$Thus, it can be seen that the value of R is removed from the equationsuch that temperature and process variations therein do not affect thevalue of the LSB. All that is necessary is to have a stable voltage,this provided by the bandgap voltage generator.

Referring now to FIGS. 10 and 10 a, there is illustrated a logic diagramfor a comparator bank, each comparator bank representing each of thecomparators 740. This comparator string is a differential comparatorhaving a positive input and a negative input. The positive input isconnected to the positive input voltage on the node 742 which isconnected to the voltage V_(IN+). The other input is connected to a node1002 which is the tap voltage V_(TAP), this reference input to thecomparator. There is provided a first comparator 1004 having a referencevoltage input on node 1006 and a primary input on a node 1008. Node 1002is connected to one side of a switch 1010, the other side thereofconnected to node 1006. Similarly, the node 742 is connected through oneside of a switch 1012, the other side thereof connected to node 1008.Node 1002 is also connected to one side of two switches 1014 and 1016,the other sides thereof connected to the nodes 1008 and 1006,respectively. Switches 1010 and 1012 are controlled by the clock signalΦ1 and the switches 1014 and 1016 are controlled by the clock signal Φ2.

The output of comparator 1004 is provided on differential outputs 1020and 1022. Output 1020 is connected to one side of a sample capacitor1024 and the node 1022 is connected to one side of a sample capacitor1026, both having a value of “C.” The other side of the capacitor 1024is connected to a node 1028, which comprises one input of a secondcomparator 1030. The other side of capacitor 1026 is connected to a node1032, which is connected to the other input of the comparator 1030, thecomparator 1030 being a differential input comparator. Node 1028 isconnected to one side of a switch 1034, and the other side thereof isconnected to a differential output node 1036 of comparator 1030.Similarly, node 1032 is connected to one side of a switch 1038, theother side thereof connected to a second differential output node 1040of differential comparator 1030. Nodes 1036 and 1040 are connected tothe differential inputs of a reconfigurable latch 1042. Switches 1034and 1038 are controlled by a clock signal Φ1′. The reconfigurable latch1042 is controlled by a clock signal Φ3. The reconfigurable latch 1042is operable to provide a latched output on differential outputs 1044 and1046 for input to the dynamic latch 1048, which is controlled by a clocksignal Φ4. This provides a latched output for input to a T-latch 1046,which is clocked by a clock signal to provide a data output, this beingthe output of the overall comparator 740.

Referring now to FIG. 11, there are illustrated timing diagrams for theclock signals associated with the embodiment of FIG. 10. The operationof the comparator bank will be described with reference to these clocksignals. When Φ1 goes high, as denoted by an edge 1102, the switches1012 and 1010 will close, resulting in the output of the respectivevoltage on the respective nodes 1020 and 1022. Shortly thereafter, theclock signal Φ1′ will go high at an edge 1104. This will result inswitches 1034 and 1038 closing, thus reducing the gain of the comparator1030 such that the voltage on nodes 1036 and 1040 is substantially thesame. At this time, the switches 1014 and 1016 are open, since the clockΦ2 is low at this time. This is the sampling operation. Thereafter, Φ1goes low at an edge 1106 and Φ2 goes high at an edge 1108, thus openingswitches 742 and 1010 and closing switches 1014 and 1016. This, ineffect, disposes the nodes 1020 and 1022 at the same voltage orsubstantially the same voltage, thus “boosting” the other side ofcapacitors 1024 and 1026 to the voltages that were previously on thenodes 1020 and 1022. In general, the voltage on the input to thecomparator 1004 on nodes 1008, 1006 comprises the difference voltageV_(IN+)−V_(TAP). The output voltage of the comparator 1004 will have anoffset voltage V_(OS1) associated therewith. This offset voltage anddifference voltage will be multiplied by the gain of comparator 1004, again A₁. Therefore, the output voltage on nodes 1020 and 1022 will beA₁(V_(IN+)−V_(TAP)+V_(OS1)). When Φ2 goes high at 1108, this representsthe “hold” operation. Therefore, this represents a sample and holdoperation. However, when switches 1014 and 1016 are closed, the voltageacross nodes 1020 and 1022 is V_(OS1) and, therefore, the voltage acrossnodes 1028 and 1032 will now be (V_(IN+)−V_(TAP)), such that the offsetvoltage associated with the comparator 1004 is effectively removed inthe hold operation.

It can further be seen that the capacitors 1024 and 1026 are isolatedfrom nodes 742 and 1002. Thus, the analog input voltage that is input onnode 742 will not be required to drive a large capacitance. Theamplifier 732 isolates the negative input voltage on node 730 from node734 and from all the subsequent tap voltages. However, the input voltageon node 742 is required to drive the inputs of each of the multiplecomparators 740. The sampling operation requires a larger capacitancefor the purpose of holding the charge for a predetermined amount oftime. Since this larger capacitor is disposed on the opposite side ofcomparator 1004, it can be seen that the need for driving a very largecapacitance and holding the voltage on that large capacitance isreduced, as the charge driven to the capacitor is driven from internalcircuitry to the comparator 1004, as opposed to a driving circuitassociated with the node 742. Thus, the drive of the sampling capacitorsis distributed among all of the comparators 740.

Referring now to FIG. 12, there is illustrated a schematic diagram ofthe transistors 718 and 726. The transistor 718 is comprised of fourbinary weighted transistors 1202, 1204, 1206 and 1208, each of thesebeing a p-channel transistor having the source/drain path thereofconnected on one side thereof to the supply voltage V_(DD). The otherside of the source/drain path thereof is connected to the node 720. Thegate of transistor 1202 is connected through the source/drain path of ap-channel transistor 1210 to node 714, the gate thereof connected to bitb0-Bar. The gate of transistor 1204 is connected to node 714 through thesource/drain path of a p-channel transistor 1212, the gate thereofconnected to bit b1-Bar. The gate of transistor 1206 is connected tonode 714 through the source/drain path of a p-channel transistor 1214,the gate thereof connected to bit b2-Bar. The gate of transistor 1208 isconnected to node 714 through the source/drain path of a p-channeltransistor 1216, the gate thereof connected to bit b3-Bar. Therefore,when the respective bits are a logic “high,” then the respective gatetransistors 1210-1216 will connect the gate of the respectivetransistors 1202-1208 to node 714. Transistors 1202-1208 are binaryweighted in size. The transistor 1202 has a size of, for referencepurposes, 1X, transistor 1204 has a size of 2X, transistor 1206 has asize of 4X and transistor 1208 has a size of 8X. Therefore, the amountof current that will flow through the transistors is correspondinglylarger. This provides the binary weighting, a fairly conventionalweighted current scheme.

When the transistors 1202-1208 are deselected, their gates will bepulled high. A pull-up p-channel transistor 1220 has the source/drainpath thereof connected between the gate of transistor 1202 and a supplyvoltage V_(DD), and the gate thereof connected to bit b0. A pull-upp-channel transistor 1222 has the source/drain path thereof connectedbetween V_(DD) and the gate of transistor 1204 and the gate thereofconnected to bit b1. A pull-up p-channel transistor 1224 has thesource/drain path thereof connected between V_(DD) and the gate oftransistor 1206 and the gate thereof connected to bit b2. A pull-upp-channel transistor 1226 has the source/drain path thereof connectedbetween V_(DD) and the gate of transistor 1208 and the gate thereofconnected to bit b3.

The transistor 726 is comprised of four n-channel transistors 1230,1232, 1234 and 1236 having the source/drain paths thereof connectedbetween node 724 and ground and sized in a binary weighted mannersimilar to transistors 1202-1208, such that they are respectivelyidentical thereto in size. The gate of transistor 1230 is connected tonode 728 through an n-channel transistor 1238, the gate thereofconnected to bit b0. The gate of transistor 1232 is connected through ann-channel gate transistor 1240 to node 728, the gate thereof connectedto bit b1. The gate of transistors 1234 is connected through ann-channel gate transistor 1242 to node 728, the gate thereof connectedto bit b2. The gate of transistor 1236 is connected through an n-channelgate transistor 1244 to node 728, the gate thereof connected to the bitb3. Thus, by selecting the ones of the gated transistors 1238-1244, thebinary weighted transistors 1230-1236 can be selectively connectedbetween node 724 and ground. When not selected, the gates thereof arepulled low through the source/drain paths of pull-down n-channeltransistors 1246, 1248, 1250 and 1252, respectively. The gates oftransistors 1246-1252 are connected to bits b0-Bar, b1-Bar, b2-Bar andb3-Bar, respectively.

Referring now to FIG. 13, there is illustrated a schematic diagram ofthe comparator 1004. This is a differential input comparator that iscomprised of two differential input n-channel transistors 1302 and 1304having the sources thereof connected in a common source configuration toa common source node 1306. Node 1306 is connected through thesource/drain path of an n-channel transistor 1305 to ground, the gatethereof connected to a bias voltage on a node 1308. A diode connectedn-channel transistor 1310 has the source/drain path thereof connectedbetween node 1308 and ground and the gate thereof connected to node1308. This provides the bias for the node 1306 for the transistor 1305.The drain of transistor 1302 is connected to a negative output node 1312and the drain of transistor 1304 is connected to a node 1314, thepositive output node. A cross coupled p-channel transistor paircomprised of a p-channel transistor 1316 connected between V_(DD) andnode 1312 at a p-channel transistor 1318 connected between V_(DD) andnode 1314 is configured such that the gate of transistor 1316 isconnected to the opposite node, node 1314, and the gate of transistor1318 is connected to the opposite node, node 1312. A diode connectedp-channel transistor 1320 is connected between V_(DD) and node 1312, thegate thereof connected to node 1312. A diode connected p-channeltransistor 1324 is connected between V_(DD) and node 1314, the gatethereof connected to node 1314. The gate of transistor 1302 is thepositive input and the gate of transistor 1304 is the negative input.

Referring now to FIG. 14, there is illustrated a schematic diagram ofthe comparator 1030. This is a differential input comparator that iscomprised of two differential input n-channel transistors 1402 and 1404having the sources thereof connected in a common source configuration toa common source node 1406. Node 1406 is connected through thesource/drain path of an n-channel transistor 1405 to ground, the gatethereof connected to a bias voltage on a node 1408. A diode connectedn-channel transistor 1410 has the source/drain path thereof connectedbetween node 1408 and ground and the gate thereof connected to node1408. This provides the bias for the node 1406 for the transistor 1405.The drain of transistor 1402 is connected to a negative output node 1412and the drain of transistor 1404 is connected to a node 1414, thepositive output node. A cross coupled p-channel transistor paircomprised of a p-channel transistor 1416 connected between V_(DD) andnode 1412 and a p-channel transistor 1418 connected between V_(DD) andnode 1414 is configured such that the gate of transistor 1416 isconnected to the opposite node, node 1414, and the gate of transistor1418 is connected to the opposite node, node 1412. A diode connectedp-channel transistor 1420 is connected between V_(DD) and node 1412, thegate thereof connected to node 1412. A diode connected p-channeltransistor 1424 is connected between V_(DD) and node 1414, the gatethereof connected to node 1414. The gate of transistor 1402 is apositive input and the gate of transistor 1404 is the negative input.This is a conventional design.

A p-channel transistor 1440 that has the source/drain path thereofconnected between nodes 1412 and 1414 and provides a short circuit for ashort duration of time prior to the leading edge of Φ1′ to preventkickback. The gate of transistor 1440 is connected to a clock signal Φ1_(pre), such that, when activated, the gain of the comparator stage 1030is substantially reduced. This clock signal is not shown in FIG. 11.

Referring now to FIG. 15, there is illustrated aschematic diagram of thereconfigurable latch 1042. This latch has two modes of operation. In thefirst mode, the gain is set at a relatively low gain and, in a secondmode, the gain is increased substantially. The input is provided by acommon source pair of differential input n-channel transistors 1502 and1504, having the source thereof connected to a common source node 1506.The n-channel transistor 1510 is connected between node 1506 and groundwith the gate thereof connected to a bias voltage on a node 1508. Thedrain of transistor 1502 is connected to a negative output node 1512 andthe drain of transistor 1504 is connected to a node 1514, the positiveoutput node. A cross-coupled pair of p-channel transistors 1516 and 1518is provided, with the source/drain path of transistor 1516 connectedbetween V_(DD) and node 15 a12nd the source/drain path of transistor1518 connected between V_(DD) and node 1514. The gate of transistor 1516is connected to node 1514 and the gate of transistor 1518 is connectedto node 1512. A p-channel transistor 1520 has the source/drain paththereof connected between nodes 1514 and 1512 and the gate thereofconnected to a node 1524. A diode connected p-channel transistor 1526 isconnected between V_(DD) and a node 1528 (p-channel), the gate thereofconnected to node 1528. A second diode connected p-channel transistor1530 is connected between node 1528 and node 1524, the gate thereofconnected to node 1524. An n-channel transistor 1532 is connectedbetween node 1524 and ground, the gate thereof connected to the biasvoltage on node 1508. A p-channel transistor 1534 has the source/drainpath thereof connected V_(DD) and node 1524, the gate thereof connectedto the clock signal Φ3. In general, the transistor 1520 is operated inthe triode region and, therefore, when turned on, constitutes aresistor. The input impedance looking into the source of transistor 1516and into the source of transistor 1518 is equal to −1/g_(m). Whentransistor 1520 is turned on, it provides a resistance, R₁₅₂₀, that isdisposed in parallel with this impedance. Initially, this is a negativeimpedance until a transistor is turned on, at which time it is impedanceabove zero, which, when turned on, results in a relatively low gain.When turned off, the gain goes high. Thus, when Φ3 goes high, node 1524is biased to place the transistor 1520 in the triode region. This occursat an edge 910 on the waveform Φ3 in FIG. 11. This occurs prior to theswitches 1014 and 1016 closing in response to Φ2 going high at the edge1108. Thus, prior to the sample operation, the latch 1042 is configuredfor a low gain operation. When Φ2 goes high at edge 1108, thereconfigurable latch 1042 will evaluate the difference voltage at thegates of transistors 1502 and 1504 which will result in a differencevoltage generated across the output nodes 1512 and 1514 with a gain oftwo. When Φ3 goes low at an edge 1112, this value will be latched on theoutputs.

Referring now to FIG. 16, there is illustrated a plot of gain of thereconfigurable latch when Φ3 is high. It can be seen that the gainvaries from a value of 4.5 at a substantially zero voltage input to avalue of 1.5 at a voltage of 100 millivolts and a voltage of 1.0 at avalue of 200 millivolts on the input.

Referring now to FIG. 17, there is illustrated a schematic diagram ofthe dynamic latch 1048. There are provided two n-channel gatetransistors 1702 and 1704 for connecting the positive and negativeinputs associated therewith to respective nodes 1706 and 1708, the gatesof transistors 1702 and 1704 gated by the Φ2 clock signal. Two commonsource n-channel transistors 1710 and 1712 have the sources thereofconnected to a common source node 1714 and the drains thereof connectedrespectively to nodes 1706 and 1708. An n-channel transistor 1716 isconnected between node 1714 and ground and controlled by the Φ4 clocksignal. Therefore, the sources of transistors 1710 and 1712 will beconnected to ground when Φ4 is a logic “high.” Node 1706 is associatedwith a positive output and node 1708 is associated with a negativeoutput. Two cross-coupled p-channel transistors 1720 and 1722 areprovided, transistor 1720 connected between a node 1724 and node 1706and transistor 1722 connected between node 1724 and node 1708. The gateof transistor 1720 is connected to node 1708 and the gate of transistor1722 is connected to node 1706. A p-channel gate transistor 1726 isprovided for connection between V_(DD) and node 1724 and the gatethereof connected to the clock signal Φ4-Bar. Thus, when transistor 1726is turned on, node 1724 is connected to V_(DD).

In operation, when the clock signal Φ2 goes high, the differentialoutput of the reconfigurable latch is connected to nodes 1706 and 1708.However, this latch is essentially powered down until the evaluationphase is complete and Φ4 goes high at an edge 114, the same time that Φ2goes low at a negative falling edge 116. Thus, the output of thereconfigurable latch which is provided at the falling edge of Φ3,falling edge 112, will be disposed on nodes 1706, and 1708 while thelatch 1048 is powered down. When transistors 1702 and 1704 are turnedoff, then the voltage on nodes 1706 and 1708 is “latched” into the latch1048 by turning on transistors 1716 and 1726. This provides an output tothe transmit latch 846.

Digital Compensator

Referring now to FIG. 18, there is illustrated a simplified diagrammaticview of the digital controller and the digital compensator 412. The PIDblock 540 is comprised of three paths that are summed with a summingjunction 1802. The first path provides a proportional relationship witha block 1804, the second path provides an integration function with anintegration block 1806 and the third block provides a differentiationpath with a block 1808. As noted hereinabove, this is referred as a PIDcontroller. The proportional block 1804 has a steady state proportionalgain, K_(p), and provides zero phase lag. The integral path andintegration block 1806 has an integral gain, K_(i), which generallyreduces the steady state error. There is some phase lag associated withthis. The differential path associated with the differentiation block1808 has a derivative gain, K_(d), which provides some phase lead byanticipating future operations. Thus, the overall PID block 540 providesphase compensation for the overall control loop.

The output of the summing junction 1802 is input to, as describedhereinabove, either a low pass filter 550 or a sinc filter 552. The lowpass filter 550 is comprised of a block 1810 that has associatedtherewith a low pass filter frequency response with two poles. This ispassed through an amplification stage 1812 with another coefficientassociated with the amplification, this being the coefficient that iscontrolled by the microcontroller 440. Thus, there are threecoefficients, a₁, a₂ and a₃ that control the operation of the low passfilter function, these being the coefficients of the low pass filter.The sinc filter 552 is basically comprised of a summing block or anaccumulation block 1814, which is operable to sum over a range of delayvalues, this being a decimation type sinc filter. A gain factor isprovided by an amplification stage 1816 which has a coefficient a₀associated therewith. This a₀ will set the position of the sinc filternotch, as will be described hereinbelow. A multiplexer 1818 is operableto select between the output of amplification stage 1812 and theamplification stage 1816 for input to the DPWM 406.

Referring now to FIG. 19, there is illustrated a more detailed blockdiagram of the PID 540 and the low pass filter 550 and the sinc filter552. The proportional path of the block 1804 has a gain stage 1902associated therewith with the gain factor K_(p). This is controlled bythe PID control block 542. The integral block has a gain block 1904associated therewith with the integral gain factor K_(i). The output ofthis is passed through a transfer function 1/(1−z⁻¹) in a block 1906.The output of this block is input to the summing junction 1802. Theintegration path and the block 1808 are comprised of a gain block 1908with a differential gain K_(d). The output of this gain block 1908 isinput to a delay block 1910 to provide the delay (1−z⁻¹). The output ofblock 1910 is input to the summing junction 1802. Additionally, there isprovided a multiplexer 1970 having one input thereof connected to theinput 1901 and the other input connected to a digital word with a valueof “0.” The output of the multiplexer 1970 is input to the input of thegain block 1904. In that error condition, the “0” value can be selectedsuch that the integration path is on hold. This will be described inmore detail hereinbelow.

The low pass filter is configured with an input summing junction 1912,the output thereof connected to a delay block 1914 with a delay of z⁻¹.The output of delay block 1914 is connected to a node 1916, which node1916 has a signal associated therewith fed back through a coefficientblock 1918 with a coefficient a₂, the output thereof input to thesumming block 1912. Node 1916 is also input to one input of a summingjunction 1918, the output thereof connected to the input of acoefficient block 1920, the output thereof providing the output of thelow pass filter on a node 1922. The input to delay block 1914 is alsoinput to summing junction 1912. Node 1916 is input through a delay block1924 with a delay z⁻¹, the output thereof input through a coefficientblock 1926 with a coefficient a₁ to another input of the summingjunction 1912. The low pass filter control block 558 sets thecoefficients a₁, a₂ and a₃. In general, this is a Butterworthconfiguration low pass filter, a fairly conventional digital filter.

The sinc filter is comprised of an input summing junction 1930, theoutput thereof input through a delay block 1932 with a delay of z⁻¹, theoutput thereof input to a coefficient block 1934, the output thereofproviding the output of the sinc filter 552, coefficient block 1934having the coefficient a₀ associated therewith, this coefficientprovided by the sinc control block 556. The output of delay block 1932is also fed back to the input of summing junction 1930 to provide theaccumulation operation. This delay block 1932 has a reset inputassociated therewith which is reset at a predetermined time. As notedhereinabove, this is a decimation type sinc filter. The output of boththe low pass filter and the sinc filter are input to respective inputsof the multiplexer 554. This provides the u(n) error signal. The lowpass filter or the sinc filter can be selected, depending upon theparticular application and the desire of the applications engineer.

Referring now to FIGS. 20 a and 20 b, the frequency response of the PID540 will be described. First, the mathematics associated with the PIDwill be set forth as follows:

$\begin{matrix}{{H\mspace{11mu}(z)_{PID}} = {K_{p} + \frac{K_{i}}{1 - z^{- 1}} + {K_{d}\left( {1 - z^{- 1}} \right)}}} \\{= \frac{{K_{p}\left( {1 - z^{- 1}} \right)} + K_{i} + {K_{d}\left( {1 - z^{- 1}} \right)}^{2}}{1 - z^{- 1}}} \\{= \frac{\left( {K_{p} + K_{i} + K_{d}} \right) + {\left( {{- K_{p}} - {2K_{d}}} \right)\; z^{- 1}} + z^{- 2}}{1 - z^{- 1}}}\end{matrix}$It can be seen from the above equations that there is a single pole atDC and that there are two zeros. Further, it can be seen that the valueof the zeros is the function of the constants K_(p), K_(i) and K_(d). Byselecting these constants, the value of the zeros can be varied.

Referring now to FIGS. 20 a and 20 b, there is illustrated frequency andphase plots for the response over frequency of the PID. It can be seenthat there is a single pole at DC and the response will roll off untilthe first zero, at which time the response will flatten out until thesecond zero. At the second zero, the response changes in a positivemanner, this due primarily to the differentiator term. However, it canbe seen that without some type of filtering, the gain at highfrequencies will be fairly high. This is the difference between adigital controller and an analog controller wherein the analogcontroller has an inherent low pass filter at the higher frequencies. Itcan be seen that the phase also exhibits a similar property wherein thephase is initially 90° and falls slightly to the first zero where itgoes positive and then at the second zero continues to increase. At highfrequencies, the phase is significantly leading in nature. With the useof a low pass filter, as set forth in FIG. 21, the high frequencyportion of the PID response can be controlled. However, the cornerfrequency of the low pass filter cannot be too low or the phaseassociated therewith will cause instability in the loop. Typically, theswitching frequency is around 500 KHz. It will be desirable to filterany noise associated with the switching frequency and, therefore, itwill be desirable from a filtering standpoint to move the cornerfrequency of the low pass filter at or below this frequency. However,this would cause significant phase instability in the control loop. Thisis where the sinc filter will be beneficial. The sinc filter, with theresponse shown in FIG. 22, results in a plurality of “notches” atmultiples of the sampling frequency, such that a notch can be placed atthe switching frequency of the power supply.

As noted hereinabove, the sinc filter is a decimation type filter. Thedecimation ratio is defined as the ratio of the sampling frequency ofthe controller divided by the switching frequency of the power supply,the desired notch. If the sampling frequency f_(s), is set at 10 MHz andthe switching frequency of the power supply, f_(sw), is equal to 500kHz, that summation ratio would be equal to f_(s)/f_(sw), which resultsin zeros at integer multiples of the switching frequency. This is equalto (10×10⁶)/(500×10³), which results in a decimation ratio of 20.Therefore, a notch would exist at 500 kHz, 1 MHz, 1.5 MHz and finally at5 MHz, f_(s)/2. Therefore, the accumulator would accumulate 20 samplesand then be reset, at which time it would provide an output.

Referring now to FIGS. 23 a and 23 b, there is illustrated a moredetailed implementational diagram of the digital compensator 412. Theinput 1901 of the PID is input along three paths, as noted hereinabove.The proportional path utilizes a multiplier 2302 having one inputconnected to the node 1901 and the other input thereof for receiving thedigital value of K_(p) and providing on the output the result for inputto a first summing junction 2304. The integral path has a multiplier2306 associated therewith having one input thereof connected to theinput 1901 and the other input thereof for multiplication with theoutput of an AND gate 2308. One input of the AND gate is connectedthrough an inverter node to an integrate hold enable signal, INTHLDEN,and the other input thereof connected to the K_(i) integral constant.The output of multiplier 2306 is fed to the input of a summing junction2308 for summing with the output of a feedback delay block 2310 which isoperable to feedback the output from a node 2312. The output of thesumming junction 2308 is passed through a saturation block 2314 to anode 2312. Node 2312 is input to the other input of the summing junction2304. The output of summing junction 2304 is input to a summing junction2316. The differentiator block has a summing node 2318 for receiving onthe one input thereof the value on the node 1901 and on the other inputthereof the value on node 1901 delayed by delay block 2320, this inputto a negative input such that the block 2318 is a difference block. Theoutput of the difference block 2318 is input to a multiplication block2322 for multiplication of the output of the summing block 2318 with theconstant case K_(d). The output of multiplication block 2322 is input tothe summing block 2316. The summing block 2316 is input to a summingblock 2324, this operable to receive on the input thereof a programmabledither signal, generated by a programmable dither register 2326. Bychanging the value of this programmable dither, the value output by thesumming junction 2316 can be varied.

The output of the summing junction 2324 comprises the PID output. Thisis input to the two filters. The low pass filter is configured with anAND gate 2330, one input connected to the PID output and the other inputthereof connected to the filter select signal, FILTERSEL-EAR. The outputof the enable gate 2330 is input to a summing junction 2332. The outputof summing junction 2332 is input to the input of a summing junction2334, the output thereof connected through a saturate block 2336 to anode 2338 for input to a delay block 2340, the output thereof connectedto a node 2342. Node 2342 is input to one input of a multiplicationblock 2344, the other input thereof connected to the coefficient a, formultiplication therewith. The output of multiplication block 2344 ispassed through a truncation block 2346 to truncate the value outputtherefrom for input to the other input of the summing junction 2334 on anegative input thereof to provide a subtraction operation with thesumming junction 2334. The output of node 2342 is also input through adelay block 2348 to the input of a multiplication block 2350 formultiplication with the a coefficient. The output of multiplicationblock 2350 is truncated with a truncation block 2352 for input to anegative input on the summing junction 2332 such that a subtractionoperation is performed by the summing junction 2332. A summing junction2358 is operable to the sum of the output of node 2342 and the output ofnode 2338, the output thereof input to a multiplication block 2360 formultiplication with the a₃ coefficient. The output of multiplicationblock 2360 is input to a block 2362 for saturation of truncation andthen to the input of the multiplexer 554.

The sinc filter is facilitated with an input selection AND gate 2364having one input connected to the PID output and the other inputconnected to the filter select signal, FILTERSEL. The output of the gate2364, the enable gate, is input to one input of a summing junction 2366,the output thereof connected through a saturate block 2368 to a node2370. Node 2370 is connected through a delay block 2372 to an input ofan AND gate 2374. The output of AND gate 2374 is input to the otherinput of the summing junction 2366. Node 2370 is also input to amultiplication block 2376 for multiplication with the sinc filtercoefficient, a₀, the output thereof connected to a saturate andtruncation block 2378 for output to the other input of the multiplexer554.

When the sinc filter is selected, a different clock signal is utilizedfor delaying the output. A delay 2380 is provided on the output of themultiplexer 554. A multiplexer 2382 selects the control signal for thedelay 2380 to adjust the delay thereof. This either can be the raw clocksignal or the raw clock signal divided by a factor of “N,” with a divideblock 2384. The clock signal is input to one input of the multiplexer2380 and to the other input thereof through the divide block 2384 toprovide the divide down clock signal. The divide down clock signal alsoprovides the second input to the enable gate 2374 through inverting nodethereon. Thus, the divide ratio provides the “reset” for theaccumulation operation, the accumulation operation operating at thefilter clock rate. The divide down “N” ratio sets the number ofaccumulations that will be allowed to occur before the reset, at whichtime the data output will be provided.

Referring now to FIG. 24, there is illustrated a Bode plot of thedigital compensator with a low pass filter. It can be seen that, at DC,there is a pole and the first zero of the PID occurs at Fz1 and thesecond zero occurs at Fz2. The response will increase at the second zerountil the first pole of the low pass filter occurs, at Fp1, and thesecond pole occurs later at a pole Fp2. Thus, it can be seen that bymoving the corner frequency of the low pass filter out from theswitching frequency and the zeros of the PID, there will be someincrease in the signal output by the PID. Of course, the two zeros ofthe PID could be identical and the two poles of the low pass filtercould be closer together.

Referring now to FIG. 25, there is illustrated a frequency plot of thesinc filter operation in the frequency domain. It can be seen that, inthis embodiment set forth hereinabove with respect to the example wherethe sampling frequency of the filter is 10 MHz and the switchingfrequency of the power supply is 500 kHz, there will be a notch 2502placed proximate to the 500 kHz switching frequency. It is noted thatthis notch is programmable to the use of the coefficients utilized torealize the sinc filter, the decimation ratio, the sampling frequencyand the switching frequency. By adjusting these values, the notch can beprogrammed for placement at the switching frequency of the power supply.This will result in a very quiet power supply, such that the switchingfrequency is effectively filtered out of the control loop.

Voltage Positioning

Referring now to FIG. 26, there is illustrated a prior art voltage plotof the voltage output in the presence of positive and negativetransients. The power supply is typically given some type ofspecification for the regulation, i.e., the regulation must be withinpredetermined limits. There is a high limit and a low limit. The reasonthat the voltage may go outside of the limits is due to ripple or due totransient responses. The ripple is typically very tightly controlled.However, transient current surges can cause the voltage to increase ordecrease. In FIG. 26, there are illustrated a positive transient and anegative transient. A positive transient will occur whenever a load isquickly removed from the output of the power supply and a negativetransient will occur when a load is applied. When a load is applied, forexample, there will be a large inrush of current. This current will havea tendency to pull the power supply voltage low and out of regulationshortly until it can be brought back into regulation. However, it may bethat the current rush will pull the voltage down below a lower limit,thus falling outside of the specification. The way the prior art systemshave accommodated this transient is to provide for a larger capacitor onthe output node. This larger capacitor will tend to reduce the effect ofthe transient and maintain it within the limit. The problem with largecapacitors is that they are expensive and large. There are two types ofcapacitors that can be utilized, ceramic capacitors or electrolyticcapacitors. The ceramic capacitors have a relatively small equivalentseries resistance (ESR), but they do not accommodate large capacitorvalues at economic costs. A typical value of a capacitor to accommodatetransients would be 100 microfarads. For this size of a capacitor, asingle discrete capacitor would typically utilize an electrolyticcapacitor. However, these electrolytic capacitors have high ESRs. Forany inductor current ripple, there would be a commensurate amount ofpower dissipated in the ESR of the capacitor. For DC voltages, therewould be no dissipation, but, for even a small ripple, there would besome heating of the capacitor. This heating could cause failure of thecapacitor, which is why ceramic capacitors are favored. Thus, eventhough the ceramic capacitor has a relatively small value, power supplymanufacturers utilize a plurality of the power supply capacitorsdisposed in parallel. Thus, for large capacitors, there can be a largepart count and, therefore, it is desirable to reduce this part count.

In the present disclosed embodiment, it is possible through the controlof the reference DAC 506 that is part of the reference generator 410, tobe controlled to reposition the set point for the reference input to theADC 408. For situations where low current is present, well below therated current of the power supply, it is anticipated that any transientwould be a negative transient due to a sub increase in the load. Thus,the set point is positioned higher than median voltage and closer to thehigher limit than the low limit. Thus, when a transient occurs, it hasthe full range between the high and low limit or substantially the fullrange, within which to pull the voltage down on the output of the powersupply. This is illustrated in FIG. 27 a, wherein the regulated DCvoltage is disposed proximate the high limit. When high current ispresent, the set point is disposed proximate to the lower limit of thepower supply specifications. When the load is removed, which would beexpected, then a positive transient would occur and, with the embodimentillustrated in FIG. 27 b, the transient can have a magnitude that hisapproximately equal to the difference between the high and low limits inthe specification.

In order to appropriately set the value output by the reference DAC 506,it is necessary to determine the current level and then set thereference voltage level accordingly. FIGS. 28 a and 28 b illustrate thisaspect. For low currents, the voltage is positioned proximate the highlimit and for the high currents, the voltage is positioned proximate thelower limit. Illustrated in FIG. 28 a is the current and 28 b is theoutput voltage position, i.e., the set point. Superimposed on thecurrent in phantom at the low current level is a current transient 2802.This results in a transient in the voltage in the transient 2804 involtage that is negative going. However, since the voltage is positionedproximate the high limit, this transient has more room than if it weredisposed at the midpoint between the high and low limits. Verysimilarly, when the current is high, there is illustrated a negativecurrent transient 2806 in phantom. This would result in the regulatedvoltage experiencing a high voltage transient 2808. It can be seen that,since the voltage is repositioned for the higher current, that more roomis allowed for the voltage mediation. The result of utilizing thevoltage positioning is that a smaller capacitor can be utilized on theoutput, which can significantly reduce the part count.

Referring now to FIG. 29, there is illustrated a flow chart for thevoltage positioning operation. This is initiated at a start block 2902and the proceeds to a block 2904 to sense the current. The current issensed with a Hall effect sensor 460 which generates a voltage outputproportional to the current, these being conventional devices. Thiscurrent is sensed and input to one of analog inputs to the chip andconverted to a digital voltage with the SAR ADC converter 660. This isstored in the associated SFR/LIM register and can be examined by thecore processor 602. Of course, any time the current exceeds the internallimit, this is accommodated by other circuitry. Once the current issensed, then the value of V_(REF) output by the reference DAC 506 isthen set. This is set in accordance with a look-up table that can bestored in the memory or any other manner to set the values of the steps,including a simple algorithm. It could be that the voltage is positionedat three points, one when the voltage is above a threshold, one when itis below a threshold and one when it is between the thresholds. However,even finer graduations could be facilitated through the use of a look-uptable.

Referring now to FIG. 30, there is illustrated a block diagram of amethod for determining the total current. Hall sensors are fairlyexpensive and, therefore, a different technique is disclosed formeasuring the currents and determining the change to the referencevoltage to be made. As noted hereinabove with respect to FIG. 1, thereis provided a half bridge power supply section 3002, which includes onthe output side a series inductor 3004. Associated with this seriesinductor 3004 is an internal resistance 3006 with a resistive value. Thecurrent through the resistor 3006 is determined with a current detector3008, which will be described in more detail hereinbelow. This providesthe inductor current to the output node 3010 wherein the output voltageV_(O) is provided. The output current, I_(OUT), is output therefrom.Disposed between this output node and ground is the load capacitor 3012,C_(OUT). This has associated therewith an internal resistance 3040. Acurrent detector 3014 is disposed between the bottom plate of thecapacitor 3012 and ground. The current detected by the current source3008 is multiplied by a gain factor and input to a summing junction3016. Similarly, the current detected by the current detector 3014 ismultiplied by a factor and input to the summing junction 3016. Ingeneral, the ratio between the two internal resistors 3006 and 3014determines what the multiplication factor is in both of the currentdetect legs. Additionally, the output voltage V_(O) is input to thesumming junction 3016. The output of the summing junction 3016 is anintermediate voltage V_(I). The output of each of the current detectlegs after the amplification stage is provided by the followingequations:V _(I) =V _(O) +I _(L) R _(ESRZ) +I _(C) R _(ESRZ)V _(I) =V _(O) +I _(OUT) R _(ESRZ)The output of the summing junction is then input to a summing junction3018 to subtract the term I_(OUT) R_(ESR2) from the value of V_(REF) toprovide the error voltage. This is input to a control block 3020 forinput to the half bridge 3002. This is one implementation, but itindicates that the current can be determined from looking at particularvoltages associated with the operation of the inductor and thecapacitor. Once the currents are known, then a factor can be determined,such as the voltage across the resistor, and this can be utilized toperform the voltage positioning. Alternatively, the absolute value ofthe current can be determined and a look-up table utilized.

Referring now to FIG. 31, there is illustrated a schematic of thetechnique for measuring the current across the resistor withoututilizing a Hall sensor. This is facilitated by disposing a series RCnetwork between one side of the inductor 3004 and the other side of theinternal resistor 3006. This is comprised of a resistor 3102 and acapacitor 3104 labeled R_(O) and C_(O). This is a monitoring circuit.If, for example, V_(O) were equal to “0,” then the followingrelationship would exist:

$\begin{matrix}{V_{1} = {I_{OUT}\left( {{sL} + R_{ESRZ}} \right)}} \\{V_{x} = \frac{\frac{1}{C_{O}\mspace{11mu} s}}{R_{O} + \frac{1}{C_{O}\mspace{11mu} s}}} \\{= {\frac{1}{1 + {s\mspace{11mu} R_{O}\; C_{O}}}V_{1}}}\end{matrix}$By combining the last two equations, the following exists:

${{I_{OUT}\left( {{sL} + R_{ESR1}} \right)}\left( \frac{1}{1 + {s\mspace{11mu} C_{O}\; R_{O}}} \right)} = V_{s}$$I_{OUT} = {V_{x}\frac{1 + {s\mspace{11mu} R_{O}C_{O}}}{R_{ESR1} + {sL}}}$$I_{OUT} = {\frac{V_{x}}{R_{ESR1}}\left( \frac{1 + {s\mspace{11mu} R_{O}C_{O}}}{1 + \frac{sL}{R_{ESR1}}} \right)}$By matching the poles and zeros of the above function, i.e., settingR_(O)C_(O)=L/R_(ESR1) or R_(O)R_(ESR1)C₀= L, thenI_(out)=B_(x)/R_(ESR1). The output of V_(x) is then equal to I_(out)mulitiplied by the value of R_(ESR1).

Referring now to FIG. 32, there is illustrated a diagrammatic view ofthe technique for determining the current through the capacitor. Thecapacitor is illustrated with an output capacitor 3202, the large filteroutput capacitor which has the internal resistance 3204. By providing aparallel series RC component comprised of a series resistor 3206 andseries capacitor 3208, it is possible to determine at a junction betweenthe resistor 3206 and capacitor 3208 a voltage which represents thevoltage across resistor 3206. This is scaled such that the voltageacross resistor 3206 is correlated with the voltage across resistor 3204and current therefrom can be measured. The relationship is as follows:

$V_{O} = {{{- I_{C}}R_{ESR}} - \frac{I_{C}}{s\; C_{O}}}$$V_{O} = {- {I_{C}\left( \frac{1 + {s\; R_{ESR}C_{O}}}{s\; C_{O}} \right)}}$$V_{CO} = {\frac{- I_{C}}{s\; C_{O}}\left( \frac{1 + {s\; R_{ESR}C_{O}}}{1 + {s\; R_{1}C_{1}}} \right)}$R₁C₁ = R_(ESR)C_(O)  $\;{V_{CO} = \frac{I_{C}}{s\; C_{O}}}\mspace{14mu}$V_(CO) = V_(O) + I_(C) R_(ESR)Thus, it can be seen that the voltage at the junction between resistor3206 and capacitor 3208 directly relates to the current through thecapacitor 3202.

Referring now to FIG. 33, there is illustrated an embodimentillustrating the current sensing of the inductor current and thecapacitor current of the embodiments of FIGS. 31 and 32. The voltageV_(x) from the junction between resistor 3102 and capacitor 3104 isinput to one side of a summing junction 3302, the other side connectedto the output voltage V_(O). The voltage V_(CO) from the junctionbetween resistor 3206 and resistor 3208 is input to a summing junction3304. Both summing junctions 3302 and 3304 are operable to subtract thevoltage V_(CO) and V_(X) from the output voltage. The output of thesumming junction 3302 is normalized to R_(ESR) by multiplying by thefunction of R_(ESR)/R₁, where R₁ is the value of resistor 3006. Thus,this output can be summed with a summing junction 3306 with the outputof summing junction 3004 to provide the voltage V_(i), the intermediatevoltage. This represents the voltage across the resistor 3014, which canthen be utilized to determine current, as this voltage represents thecurrent I_(OUT) through a resistor of a value R_(ESR). From animplementation standpoint, the voltage V_(X) on the junction betweenresistors 3102 and 3104 is provided as an input to one of the analoginputs on the pin 652 for input to the multiplexer 656. The V_(CO)output at the junction between resistors 3206 and 3208 is also providedas analog input. All the microcontroller requires is knowledge of thevalues of the resistor 3006 and the resistor 3014 in order to determinethe current through the inductor and capacitor, respectively. Thefunctions R_(ESR)/R₁ is a constant that can be determined from knownvalues and this utilized in the microcontroller to perform theoperations of the summing junctions 3302, 3304 and 3306 and thenormalization stage.

DPWM

Referring now to FIG. 34, there is illustrated a general block diagramof the DPWM 416. As noted hereinabove, the DPWM 416 is a state machinethat is operable to generate up to six phases for use in drivingexternal drivers that will control switches on the switching powerconverter. Each of these phases will be defined by a leading edge and atrailing edge, either leading or trailing edge being rising or falling.Either the u(n) signal from the digital compensator 412 or amicrocontroller generated PID value from the register 564 is provided onthe bus 570 as an input to the DPWM 416. The DPWM 416 provides forhighly flexible operation, which is operable to accommodate variouspulse width and phase modulation schemes. Phase-to-phase timing can beprogrammed for fixed (or zero) dead time, or the microcontroller 440 candynamically control dead time during converter operation. The DPWM 416may be clocked at 200 MHz (5 nS resolution) or 50 MHz (20 nSresolution), depending on the setting associated therewith, these clocksgenerated by the PLL. It is noted that the DPWM is a state machine, suchthat, for each clock cycle, there is a result output by the statemachine, as compared to an instruction based microprocessor or a DSPsolution.

There are provided two paths from the input bus 570. The first path isassociated with a Symmetry Lock logic block 3402, which is operable tointerface with a Symmetry Lock SFR in the microcontroller 440. As notedhereinabove, there are a plurality of SFRs, some of which are notillustrated, each of these SFRs occupying a portion of the address spaceof the microcontroller 440, such that they can be written to or readfrom. The Symmetry Lock logic block 3402 is operable to latch each valueoutput by the multiplexer 566 upon receiving a Data Ready signal. Sincethe digital compensator 412 operates at a clock rate of 10 MHz with aswitching frequency of 500 kHz, for example, there will be many moresamples of u(n) during a particular switching frame than may berequired. However, u(n) can be changing and there may be modulationschemes and phase schemes that require an edge of the pulse to be sentbased upon current data. The block 3402 latches each value and, upon theoccurrence of a predetermined lock condition, the data will be “locked”into the logic block 3402. This situation occurs when, for example, thetrailing edge of PH1 requires current data to determine the positionthereof. Once the trailing edge occurs on PH1, a system may be set upthat, for example, the leading edge of PH1, a relative edge to thetrailing edge of PH1, calculates its position relative to PH1 based uponthe locked data in the logic block 3402.

There are provided two separate paths output from the logic block 3402,a first path associated with a summation block 3404 and a second pathassociated with a summation block 3406. Each of the summation blocks3404 and 3406 is operable to receive a 2's complement correction datavalue from a correction data SFR, labeled TLCD0 and TLCD1, respectively,which basically each provide an offset. There may be situations wherethe designer needs to compensate the mismatch of the components in thepower supply. As such, it may be desirable to increase or decrease thevalue of u(n). Once corrected, each of the paths flows to an associatedlimit block 3408 and 3410, respectively, which will provide a correctedu(n). The limit block 3408 is associated with the summation block 3404and receives high and low limits, TLGT0 and TLLT0, wherein the limitblock 3410 associated with the summation block 3406 receives high andlow limits TLGT1 and TLLT1, these limits associated with respectiveSFRs. By providing two correction paths for each Symmetry Lock logicblock, this allows a first edge to be defined based upon currentlychanging data and then subjected to two different correction factors andtwo different limit factors.

There is provided a second Symmetry Lock logic path associated with alogic block 3412, having associated therewith two paths associated with,in the first path, a summation block 3414 and the limit block 3416. Thesecond path has associated therewith a summation block 3418 and a limitblock 3420. These blocks have associated correction data and associatedhigh/low limit values. This will provide two additional corrected u(n)values which can both be locked.

Each of the four corrected u(n) values form the blocks 3408, 3410, 3416and 3420 are input to a timing generator 3422 which generates the phasevalues for output to a timing generator bypass logic block 3424. Thelength of a switching cycle can be defined by signal SWC_CYC and thereis also provided via control of the microcontroller 440 a start of cyclesignal DPWM_EN. The polarity of the initial pulse edge, rising orfalling, is determined by PH_POL.

Referring now to FIG. 35, there is illustrated a more detailed blockdiagram of the Symmetry Lock logic circuit. When enabled, the twoSymmetry Lock logic blocks 3402 and 3412 store the value of u(n) onceper switching cycle at a time specified by a register 3502, DPWMULOCK.The two latch u(n) values are paired with two trim and limit functions,resulting in four unique corrected u(n) functions, resulting in fourunique corrected u(n) functions that can be mapped to any of the PHnoutputs in any combination. The value of u(n) on the databus 570 isinput to a data input of two latches 3504 and 3506 at the data inputthereof. Each of the latches has a clock input. The clock input of latch3504 is connected to the output of a leading/trailing edge select block3508 which is controlled by the bit ULCK0_EDG bit of the register 3502.This is operable to select either the leading or trailing edge of one ofthe six phases PH1-PH6 that are selected by a multiplexer 3510. This iscontrolled by the first three bits of the register 3502. As noted, thelatch is operable to latch each value of the u(n) data therein. Theoutput of the latch 3504 is input to the trim and limit blocks 3512 and3514, associated with the blocks 3404, 3408, 3406 and 3410 of FIG. 34.Similarly, there is provided in the second Symmetry Lock logic path amultiplexer 3516 controlled by the bits 4-6 of register 3502 forselecting one of the six phases and inputting that to a leading/trailingedge select block 3518, the output thereof driving the clock input ofthe latch 3506. The latch 3506 is associated with two trim and limitblocks 3520 and 3522 that correspond to blocks 3414 and 3416, and blocks3418 and 3420.

The timing generator 3422 is comprised of a plurality of multiplexersand phase generators. Each path has a multiplexer 3526 associatedtherewith and a phase generator 3528, each of these being a patterngenerator. Each multiplexer 3526 is operable to receive all four of thecorrected u(n) values and, depending upon which one is mapped to theparticular phase path, input that to the associated pattern generator3528.

Referring now to FIGS. 36 a and 36 b, there is illustrated a moredetailed diagram of the trim and limit sub-system, illustrating theregisters and how they interface with various function blocks. Amultiplexer 3630 is operable to be disposed between each of the outputsof the limiters 3408, 3410, 3416 and 3420 for forcing the operand to “0”such that the duty cycle of the output PH1-PH6 will be terminated whenICYC IRQ happens. This provides protection to the system from exposureto long term over current conditions. Note that, although themultiplexer 3630 is illustrated as a single multiplexer, there isactually a separate multiplexer for each cu(n) output.

Referring now to FIG. 37, there is illustrated a more detailed blockdiagram of the pattern generator 3528 for one of the phases. Each of thephase generators is divided into two sections, one for processing theleading edge and one for processing a trailing edge. As will bedescribed hereinbelow, each phase generator is based upon a leading ortrailing edge. It is the generation and positioning of this edge that ishandled by the state machine. Each edge is associated with a specifictiming type. The timing type is an absolute time, wherein edge isdefined as one that unconditionally occurs at a specific time-tick.Relative time is associated with an edge that occurs a prescribed timeafter its reference edge transitions. For example, normally PH1 has aleading edge that occurs at time-tick #1 with a pulse width that isdefined as a finite value of the u(n). The leading edge of PH2 is arelative edge, in some power converters, wherein the leading edgethereof occurs a number of time-ticks after the falling edge of PH1.Another is hardware modulation timing, which is associated with an edgethat occurs at a time specified by the value of one of the fourcorrected u(n) modulation terms.

Each of the leading edge and trailing edge functionalities haveassociated therewith a portion of the multiplexer 3526. The leading edgeportion has a multiplexer 3702 associated therewith which is operable toselect one of the four corrected u(n) values, or a “relative” or a“absolute” input. These inputs are provided by the PHn_CNTL as oneregister for the lowermost eight bits and a ninth bit from the PHn_CNTL0register. These are SFR control registers. This provides a leading edgecontrol value for the leading edge portion. Configuration data isprovided that is the select input to the multiplexer 3702 and isprovided by the three lowermost bits of a PHn_CNTL0 SFR. The informationin these three bits is also input, along with the output of themultiplexer 3702, to a phase bit logic block 3704 that is operable tocarry out the operations associated with defining the leading edge anddefining the trailing edge. Thus, a control value is what is provided bythe multiplexer 3702. The reference phases are provided by multiplexer3706 that selects between one of the six phases as a reference phase inthe event that this is a relative edge created. This multiplexer iscontrolled by the four lowermost bits of the PHn_CNTL0 SFR.

The trailing edge is handled in a similar manner to the leading edge inthat a multiplexer 3710 is provided for receiving the four correctedu(n) values and also Relative and Absolute inputs from the PHn_CNTL3control register and the eighth bit from the PHn_CNTL2 register. Thisprovides a trailing edge control. A multiplexer 3712, similar to themultiplexer 3706, selects one of the PH1-PH6 phases as the referencephase when a relative edge is being generated, and this is controlled bythe four lowermost bits of the PHn_CNTL2 register.

The contents of the PHn_CNTL0 and CNTL2 registers is set forth in Tables1 and 2.

TABLE 1 PHn_CNTL0: Phase n Leading Edge Control Register 0 Bit 7PHnL(8): PHn Leading Edge Timing Data Bit 8 This is bit 9 (MSB) of thePHn_CNTL1 register Bit 6-4 PHnL_SEL2-0: Phase 3 Leading Edge ControlBits 000: PHn Leading Edge Timing Determined by u(n)0 001: PHn LeadingEdge Timing Determined by u(n)1 010: PHn Leading Edge Timing Determinedby u(n)2 011: PHn Leading Edge Timing Determined by u(n)3 100: PHnLeading Edge Timing is Relative to Another Timing Edge 101: PHn LeadingEdge Timing is Relative to Another Timing Edge 110: PHn Leading EdgeTiming is Relative to Another Timing Edge 111: PHn Leading Edge Timingis Absolute Bit 3 PHnL_EDGE: Relative Training Reference EdgeLeading/Trailing Edge Select  0: Relative Timing is Referenced toLeading Edge  1: Relative Timing is Referenced to Trailing Edge Bit 2-0PhnL_PHn-0: PHn Leading Edge Relative Timing Reference Edge 001: PHnLeading Edge Timing Relative to PH1 010: PHn Leading Edge TimingRelative to PH2 011: PHn Leading Edge Timing Relative to PH3 100: PHnLeading Edge Timing Relative to PH4 101: PHn Leading Edge TimingRelative to PH5 110: PHn Leading Edge Timing Relative to PH6

TABLE 2 PHn_CNTh2: Phase n Trailing Edge Control Register 2 Bit 7PHnT(8): PHn Trailing Edge Timing Data Bit 8 This is bit 9 (MSB) of thePHn_CNTL1 register Bit 6-4 PHnT_SELn-0: Phase 2 Trailing Edge ControlBits 000: PHn Trailing Edge Timing Determined by u(n)0 001: PHn TrailingEdge Timing Determined by u(n)1 010: PHn Trailing Edge Timing Determinedby u(n)2 011: PHn Trailing Edge Timing Determined by u(n)3 100: PHnTrailing Edge Timing is Relative to Another Timing Edge 101: PHnTrailing Edge Timing is Relative to Another Timing Edge 110: PHnTrailing Edge Timing is Relative to Another Timing Edge 111: PHnTrailing Edge Timing is Absolute Bit 3 PHnT_EDGE: Relative TrainingReference Edge Leading/Trailing Edge Select  0: Relative Timing isReferenced to Leading Edge  1: Relative Timing is Referenced to TrailingEdge Bit 2-0 PHnT_PHn-0: PHn Trailing Edge Relative Timing ReferenceEdge 001: PHn Trailing Edge Timing Relative to PH1 010: PHn TrailingEdge Timing Relative to PH2 011: PHn Trailing Edge Timing Relative toPH3 100: PHn Trailing Edge Timing Relative to PH4 101: PHn Trailing EdgeTiming Relative to PH5 110: PHn Trailing Edge Timing Relative to PH6

Referring now to FIG. 38, there is illustrated a diagrammatic view ofthe bypass logic block 3424. The bypass logic is operable to safeguardthe power supply system by forcing each PH output into user-defined“safe” states during supply shutdown. The embodiment of FIG. 38 showsthe bypass logic for each phase. An output multiplexer 3802 is operableto select, on one input, the output of the DPWM pattern generator 3422,the default output, or one of three static pre-defined states containedin the Software Bypass (SWBP_OUT) SFR 3806, the over-current protectionfault (OCP_OUT) SFR 3808, or the Enable (ENABX_OUT) bypass SFR 3810.Therefore, the three shutdown sources, in priority, are the enableinput, the over-current protection fault and the software bypass (notingthat the software bypass is initiated by the microcontroller 440). Boththe ENABLE input and the OCP are hardware shutdowns and are enabled bysetting a bit in the DPWMCN register to a logic “1” which bit is theHWBP_EN bit. When enabled, a supply shutdown occurs when either theenable input pin is forced to its off state or the over-currentprotection interrupt (OCPIRQ) interrupts are asserted. If both occursimultaneously, the higher priority ENABLE interrupt will prevail. Thelowest priority shutdown source is software bypass, which is invoked bythe microcontroller 440 by setting an SWBP bit in the DPWMCN register toa logic “1.” This is all facilitated with a priority encoder 3812wherein the DPWMCN SFR is connected to the highest priority inputthrough an AND gate 3814 on one input thereof, the ENABIRQ interruptconnected to the other input. The DPWMCN HWBP_EN bit is also input toone input of an AND gate 3816, which has the output thereof connected tothe second priority input, with the other input of AND gate 3816connected to the OCPIRQ interrupt. The SWBP bit of the DPWMCN registeris connected to one input of an AND gate 3818, the output thereofconnected to the lowest priority input of the encoder 3812, the otherinputs of the AND gate 3818 connected to the SWBP_PHnEN bit associatedwith the particular phase. The transition from DPWM output to any of thethree-defined states can be programmed to occur on switching frameboundaries, or instantaneously by setting the EMGY_EN bit of the DPWMCNregister to a “1.” This is input to a control input of a multiplexer3820, which selects either the direct output of the priority encoder3812 or the output of an end-of-frame synchronizer block 3824 disposedbetween the other input of multiplexer 3820 and the output of thepriority encoder 3812. The frame synchronizer block 3824 is controlledby the EOFIRQ, the end-of-frame interrupt. For this end-of-framesynchronizing, this indicates that all operations, the generation of allleading and trailing edges for all phases, will occur prior to thebypass mode. With the use of the bypass safeguards, the state of each ofthe phases can be predetermined. In this manner, it can be insured thata transistor will not be closed and power being directed toward aninductor to basically destroy the transistor or other parts of the powerconverter.

Referring now to FIG. 39, there is illustrated a timing diagramdepicting the Sync Mode of operation. This mode allows the start of eachswitching cycle to be synchronized with an external clock. The userenables sync mode by assigning the SYNC input to the port I/O pins bysetting a sync enable signal, SYNCEN, in the XBAR0 SFR and the SYNC_ENbit in the DPWMCN to a logic “1.” A logic level sync pulse is applied tothe SYNC input of the integrated circuit, the positive edge of whichtriggers (or re-triggers) the start of a new switching operation, shownin FIG. 39. It can be seen that when the SYNC input goes high, at anedge 3902, the switching cycle will terminate. The SYNC pulse mustreturn low a minimum of three clock cycles of the DPWM prior to the nextpositive transition, as illustrated in the associated detail. Theswitching cycle in execution is unconditionally terminated and the newswitching cycle initiated on the positive edge 3902 of the SYNC pulse.In non-sync mode, SWC_CYC in SFR register defines the switching cycleperiod.

Referring now to FIG. 40, there is illustrated a timing diagram for theoperation of frame skipping, with FIG. 41 illustrating a detail of thebypass operation of FIG. 38. In the illustration of FIG. 41, theSWBP_PHnEN bit of the SWBP_OUTEN SFR is input to one input of an ANDgate 4102, the other input thereof connected to the SWBP bit of theDPWMCN register. For frame skipping, even at a minimum PWM duty cycle,system losses at minimum may be insufficient to prevent V_(OUT) fromrising above its specified maximum. Frame skipping reduces the effect ofenergy transferred to the load by momentarily shutting the supply outputoff on alternate cycles. It is analogous to pure skipping, but appliesto all PH outputs. In reference to the timing diagram of FIG. 40, it canbe seen that each PHn bit has a corresponding PHn enable bit inSWBP_OUTEN and a SWBP bit in SWBP_OUT. The end-of-frame interruptinterrupts the microcontroller 440 at the end of each switching cycle.When this occurs, the microcontroller 440 will clear the SWBP bit inDPWMCN register, forcing the output multiplexer 3802 for each PH outputto pass either the DPWM output (active switching cycle), or the OFFstate contained in SWBP_OUT. Frame skipping can be configured to skipany number of cycles. Normal (continuous active frame) load resumes whenfirmware detects an increase in output loading.

Referring now to FIG. 42, there is illustrated a flow chart depictingthe operation of creating an edge in a driving pulse in the patterngenerator. Each edge is created similar to another edge by thisprocedure, such that the pattern generator is operable to operate onedges, such that all that needs to be defined by the power supplydesigner is the parameters of an edge, whether it is an absolute edge, arelative edge, how many ticks to wait before generating the edge,whether it is a leading edge or a trailing edge and whether it isfalling or rising. The program is initiated at a block 4202 wherein thereferenced edge is selected in the appropriate phase, i.e., either theleading edge or the trailing edge. The program then flows to a decisionblock 4204 to determine if it is an absolute edge. An absolute edge, asdescribed hereinabove, is one that has a defined starting point from theedge of the initiation of the frame. If so, the program flows along the“Y” path to a function block 4206 to reset the base to a value of “0.”If it is not an absolute edge, then the program flows along the “N” pathto a function block 4208 to monitor for the reference edge, i.e., todetermine when the reference edge has occurred. This program flows to adecision block 4210 to determine if it has been triggered and it willmaintain itself in a loop until such time, at which time it will flowalong the “Y” path to a function block 4212 to latch the value of theDPWM counter as the base value. This basically sets the reference edgeas the base. The program then flows to a decision block 4214 todetermine if the corrected value of u(n) is selected. This is also thepoint in the program to which the function block 4206 flows. If thecorrected u(n) is selected, the program flows along the “Y” path to afunction block 4216 to select the corrected u(n) as the operand. If not,the program flows along the “N” path to a function block 4218 to selectthe value from the SFR register as the operand, this being a fixedvalue. Both function blocks 4216 and 4218 flow to a function block 4220to set the expected value equal to the base value plus the value of theoperand. The program then flows to a function block 4222 wherein theresult is compared with the DPWM counter value. A decision block 4224determines when the expected value is greater than or equal to the DPWMcounter value, at which time it will flow along a “N” path to a functionblock 4226 to trigger the edge, i.e., create the edge. The program thenflows to a function block 4228 to monitor for the end of frame interruptand, if it occurs, the program will flow from a decision block 4230along a “Y” path to a function block 4232 in order to reset the edge andthe state machine, at which time the program will flow back to the inputof function block 4202.

Referring now to FIG. 43, there is illustrated a flow chart depictingthe operation of selecting the value of u(n) from either the PID or theSFR. The program is initiated at a block 4302 and then proceeds to ablock 4304 to monitor if the corrected u(n) value is ready for latching.As described hereinabove, the data is ready after it has been processedthrough the conversion cycle of the ADC and then passes through thedigital compensator. At the digital compensator, for example, the sincfilter may take longer to process due to the decimation aspect thereof.The program then flows to a decision block 4306 to determine if a newu(n) is ready and, if not, it loops back to function block 4304. Whenready, the program flows to a decision block 4308 to determine if theSymmetry Lock edge has been triggered such that Symmetry Lock ispresent. If so, this indicates that the new u(n) should not be processedand the program flows back to the input of the function block 4304. Ifnot, the program then flows to a function block 4310 to latch the newu(n) into the register and then to a function block 4312 to correct thevalue of u(n) by the offset to provide a corrected value thereof. Theprogram then flows to a decision block 4314 to determine if a correctedvalue of u(n) is less than the minimum limit and, if so, then it flowsto a function block 4316 to set the corrected value of u(n) to theminimum limit. If not, then the program flows to a decision block 4318to determine if the corrected value of u(n) is greater than the maximumand, if so, the program flows to a function block 4320 to set thecorrected value of u(n) to the maximum limit. If neither limit has beenbreached, the program flows to a function block 4322 to leave thecorrected value of u(n) unchanged and then to a decision block 4324 todetermine if the ICYC interrupt has occurred. If so, the program flowsto a function block 4326 to set the corrected value of u(n) to “0” and,if the interrupt has not occurred, the program flows to a function block4328 to leave the corrected value of u(n) unchanged.

Referring now to FIG. 44 a, there is more fully illustrated the overcurrent protection circuitry 4400 of the digital pulse width modulatorcircuit 416 contained within block 446 (FIG. 4). The over currentprotection circuitry 4400 has provided thereto a voltage related to theoutput current IPK of the buck converter 402. The output current IPK ismeasured via a hall sensor which provides the measured current output.The voltage related to the output current IPK is provided to thepositive input of a comparator 4402 via input line 4404. The switch 4406on the input line 4404 is associated with the leading edge blankercircuit 4408 which be more fully discussed herein below. The negativeinput of the comparator 4402 is connected to the output of a 4-bitprogrammable digital to analog controller (DAC) 4410. The 4-bitprogrammable DAC 4410 provides a voltage related to the thresholdcurrent I_(TH) to the negative input of comparator 4402. The 4-bitprogrammable DAC 4410 is programmed to provide a desired threshold by acontrol register 4412 having a control value stored therein. Thecomparator 4402 compares the provided voltage related to the outputcurrent IPK of the buck converter 402 with the programmed voltagerelated to the threshold current I_(TH) and when the voltage related toIPK exceeds the voltage related to the threshold current I_(TH), aprimary interrupt (ICYCIRQ) is generated on line 4414 from the output ofcomparator 4402. The value to which the voltage related to the I_(TH)current is programmed by the digital to analog controller 4410 is basedupon the limits of the buck converter 402 to which the DPWM isconnected. Hysteresis for the comparator 4402 is controlled fromhysteresis control values from a control register 4416. The primaryinterrupt (ICYCIRQ) is provided to a clock input of 5-bit counter 4418.The primary interrupt (ICYCIRQ) is also provided to the input of resetlogic 4420. The primary interrupt is output via line 4422 to the DPWM416, the controller 440 and to the integrator stage of the PID 540.

The 5-bit control register 4418 monitors the number of occurrences ofthe primary interrupt. The present count for the number of occurrencesis provided as an output on line 4424. The present primary interruptcount is stored within a control register 4426 called ICYC count. Thepresent ICYC count on line 4424 is compared at a comparator 4428 with anover current protection count limit provided from register 4430. The OCPcurrent limit comprises the maximum number of occurrences of primaryinterrupt ICYCIRQ in consecutive frames that may occur. The present ICYCcount from the 5-bit counter 4418 is compared with the OCP count limit,which is stored in register 4430, at comparator 4428, and if the ICYCcount from the 5-bit counter 4418 equals the OCP count limit, asecondary interrupt OCPIRQ is generated from the comparator 4428 onoutput line 4432. The secondary over current interrupt is provided tothe DPWM 416 to indicate the occurrence of a serious over currentcondition.

The primary over protection interrupt ICYCIRQ provides an indication ofover current conditions which may or may not fix themselves in a nextframe period. The occurrence of consecutive primary interrupt conditionsare monitored by the 5-bit counter 4418 such that when a predeterminednumber of primary interrupts have occurred, the secondary interruptOCPIRQ may be generated to indicate a more serious over current problemsuch as a dead short. The primary interrupt ICYCIRQ performs a number offunctions within the switch power supply device described with respectto FIG. 1. The primary interrupt ICYCIRQ is provided to the DPWM 416such that each of the switches connected to the phase outputs of theDPWM 416 are turned off. Additionally, the primary interrupt ICYCIRQ isprovided to the PID 540 to hold the integrator to prevent it fromoverloading.

Referring now to FIG. 44 b, there is illustrated the circuit forproviding the integrated hold to the PID 540. The primary interruptICYCIRQ is applied to a first input of OR gate 4470. The second input ofOR gate 4470 is connected to the integrator hold output from a latch4472. The output comprises the Q output of the latch 4472. The output ofOR gate 4470 is applied to an input of AND gate 4474. The other input ofAND gate 4474 is an inverted input of the end of frame interrupt EOFIRQ.The output of AND gate 4474 is connected to the D input of latch 4472. Aclock signal PWMCK is applied to the clock input of the latch 4472.

FIG. 44 c describes the operation of the circuit of FIG. 44 b. At step4480, the integrator hold circuit monitors for the primary interruptICYCIRQ. Inquiry step 4482 determines if the ICYCIRQ interrupt has beendetected. If not, control passes back to step 4480. Once the primaryinterrupt is detected, the integrator hold circuit is initiated at step4484. Once the integrator hold circuit has been initiated, inquiry step4486 determines if the end of frame interrupt has been received. If not,the integrator hold circuit remains active at step 4484. Once the end offrame interrupt is detected, the integrator hold circuit is released atstep 4488.

This is more fully illustrated in FIG. 45 where there is shown thepulsed output 4502 associated with PHX which could be any phase outputsof the DPWM 416, and the primary interrupt signal ICYCIRQ provided fromthe output of the comparator 4402. FIG. 45 illustrates three separateframe periods. Occurring from times T₀ to T₁ is a first frame 4506 a,from time T₁ to time T₂ is a second frame 4506 b and from time period T₂to time period T₃ is a third frame 4506 c. During time frame 4506 a, aswitch connected to the output of PHX would be turned on by the risingpulse edge 4508. Upon detection of a pulse indicating a primaryinterrupt at rising edge 4510, the switch connected with output PHXwould be turned off by the signal being driven low at 4512 by the DPWM416. Likewise, in frame 4506 b, the switch associated with DPWM outputPHX would be turned on at 4514 and turned off at 4516 responsive todetection of the primary interrupt ICYCIRQ at 4518. The turning off of aswitch in response to detection of the ICYC interrupt occurs similarlyin frame 4506 c.

If the over current condition continues over multiple frames and thesecondary interrupt OCPIRQ is generated, this signal is provided to theDPWM 416 which then has the option of immediately stopping operation ofthe DPWM upon receipt of the secondary interrupt OCPIRQ, oralternatively, may wait to cease operation of the DPWM at the end of thenext frame. Whether the DPWM ceases operation right away or at the endof the frame is programmable by the user.

Referring now back to FIG. 44 a, the reset logic 4422 is responsive tothe primary interrupt ICYCIRQ and the end of frame interrupt EOFIRQprovided from the DPWM 416 to reset the 5-bit counter to “0” when pulsesof the primary interrupt ICYCIRQ are no longer received in consecutiveframes. Thus, if the reset logic 4420 within a previous frame hasdetected occurrence of a primary interrupt ICYCIRQ, and in the nextframe, as indicated by the occurrence of the end of frame interruptEOFIRQ, there is detected no occurrence of the primary interruptICYCIRQ, the reset logic 4420 provides a signal to the reset input ofthe 5-bit counter 4418 via line 4440 to reset the 5-bit counter to “0.”The end of frame interrupt EOFIRQ is additionally provided as an inputto the 5-bit counter 4418. This enables the 5-bit counter to only counta single occurrence of the primary interrupt ICYCIRQ within a particularframe. If the 5-bit counter 4418 had already counted the occurrence of aprimary interrupt ICYCIRQ during a single frame period and receives asecond primary interrupt pulse, the counter 4418 will not count thispulse since the counter had not received an end of frame interrupt sincereceiving the last ICYCIRQ primary interrupt.

The leading edge blanker circuit 4408 mentioned herein above receives aninput from the leading edge blanker select register 4442. The leadingedge blanker select register 4442 provides a control input for actuatingor not actuating the leading edge blanker circuit 4408. The leading edgeblanker select register 4442 also provides an indication to the phaseselector 4443 of the phase output of the DPWM 416 that is to be blanked.The phase selector 4443 is connected to receive each of the PH1-PH6outputs of the DPWM 416, such that the leading edge blanker circuit mayknow when to actuate a leading edge blanker output via output 4444 toswitch 4406 corresponding to a leading edge on one of these phaseoutputs. The leading edge blanker select register 4442 also provides thelength of the blanking time of the blanking pulse. Additionally, theleading edge blanker circuit 4408 receives an input from the end offrame interrupt EOFIRQ to indicate when a frame has ended. This enablesthe leading edge blanker circuit 4408 to know when to begin looking fora next leading edge pulse. Finally, the PWMCK is a clock input clockingoperations of the leading edge blanker circuit 4408. The output of theleading edge blanker circuit 4408 is provided to switch 4406 to providean open switch condition at switch 4406 to keep the input of thecomparator 4402 from seeing a spiked current output on the IPK line.This is more fully illustrated in FIG. 46.

FIG. 46 illustrates the output of one of the phase outputs 4602 from theDPWM 416, the output current IPK 4604 and the blanking signal 4606.Within a first frame 4608, the phase output of one of the outputs of theDPWM circuit 416 goes high at 4610. This comprises the leading edge ofthis switching pulse. In response to the output 4602 going high at 4610,a current spike 4612 due to parasitic capacitance is created at thecurrent output IPK. If the voltage related to the current spike 4612were applied to the input of the comparator 4402, the comparator 4402might inadvertently register an over current condition responsive to thecurrent spike even though no over current condition actually existed. Ablanking pulse is provided from the leading edge blanker circuit 4408via the output 4444 to the blanking switch 4406 to set the switch to anopen condition to keep the comparator 4402 from monitoring the currentspike on IPK. The current blanking pulse 4614 will only open theblanking switch 4406 during the time of current spike 4612. Theremainder of the time the switch is closed enabling the comparator 4402to compare the output current to the threshold current. The operation ofthe blanking signal 4606 in the following frame 4616 occurs in a similarfashion. The phase blanked by the leading edge blanker circuit 4408 andthe length of the blanking pulse 4614 are each programmable by the userthrough the LEB select register 4442. The blanking circuit 4408 may alsodetect a falling edge signal that comprises a leading edge signal.

Referring now to FIG. 47, there is illustrated a flow diagram describingthe operation of the over current protection circuitry in the manner forgenerating both the primary interrupt ICYCIRQ and the secondaryinterrupt OCPIRQ. The leading edge blanker circuit initially monitors atstep 4702 the output current IPK. The output current IPK is compared atstep 4704 with the threshold current I_(TH) to determine whether theoutput current exceeds the threshold current. If inquiry step 4706determines that the output current does not exceed the thresholdcurrent, control passes back to monitoring step 4702.

Once the inquiry step 4706 determines that the output current hasexceeded the threshold current, a primary interrupt ICYCIRQ is generatedat step 4708. Inquiry step 4710 determines if the interrupt is occurringwithin a new frame. If not, control passes back to monitoring step 4702to continue to monitor for the occurrence of a primary interrupt in anew frame. If inquiry step 4710 determines that the primary interrupthas occurred within a new frame, the interrupt count is incremented atstep 4710.

Next, at inquiry step 4714, a determination is made if the interruptcount has reached the count limit. If not, control returns to monitoringstep 4702 to begin monitoring for a next interrupt pulse. If theinterrupt count limit has been equaled, a secondary interrupt OCPIRQ isgenerated at step 4716. The controller 440 will reset the OCPIRQ whenthe OCP condition is removed, and process flow returns to monitoringstep 4702 to continue monitoring the output current.

Referring now to FIG. 48, there is illustrated the process of operationof the reset logic 4420. The reset logic 4420 monitors at step 4802 theoccurrence of the primary interrupt from the comparator 4402. If inquirystep 4804 detects an interrupt, control passes back to monitoring step4802. If no interrupt is detected, inquiry step 4806 determines if anend of frame interrupt has been received by the reset logic 4420. If noend of frame interrupt has been received, control passes back to step4802 to continue monitoring the primary interrupt output. When inquirystep 4806 detects an occurrence of an end of frame interrupt and noprimary interrupt has been detected within that frame, the counter 4418is reset at step 4808. Control then returns to monitoring step 4802 torepeat the process.

Referring now to FIG. 49, there is illustrated the circuitry forproviding both over voltage and temperature protection for the DPWM 416contained within block 446 (FIG. 4). A number of analog signals areapplied to the input of a multiplexer 4902. These signals are providedfrom various analog outputs and include a VSENSE input sensing theoutput voltage of the switched power supply and an AIN0/VIN input whichis monitoring the input voltage of the switched power supply. Also, aTEMP signal is provided by a temperature sensor 4904 that measures thetemperature of the device. These signals are multiplexed to the output4906 of the multiplexer 4902 and provided to the input of a 12 bitanalog to digital converter (ADC) 4910. The 12 bit ADC 4910 iscontrolled from values from an ADC control register 4912. The output ofthe 12 bit ADC is a digital output which is applied to the input of aspecial function register/limit (SFRILIM) register set. There are anumber of SFR/LIM register sets associated with output of the ADC 4910.Each of the SFR/LIM register sets are associated with one of the inputanalog signals provided to the multiplexer 4902. The SFR/LIM registersets have stored therein a limit value. The SFR/LIM register setcompares a provided input from the ADC 4910 to this limit value, and ifthe limit value is exceeded, generates an associated interrupt signal atthe output of the SFR/LIM register set.

Thus, when the VSENSE signal is applied to the input of the 12 bit ADC4910, a digital VSENSE signal is applied to the input of SFR/LIMregister set 4920. The SFR/LIM register set 4920 compares the provideddigital value of VSENSE to the predetermined value stored within theregister set 4920. If the provided value exceeds the stored value, aVSENSEIRQ is generated at output 4922. If the provided value does notexceed the stored limit value in register set 4920, no VSENSEIRQ isgenerated. Likewise, if the VIN value is applied to the input of the 12bit ADC 4910, the digitized value is applied to the input of SFR/LIM4924. If the provided digital value of the VIN exceeds the stored limitvalue in the register set 4924, a AIN0/VINIRQ is generated at output4926. The remaining SFR/LIM register sets operate in a similar mannerresponsive to a digital input that is compared to a limit value storedwithin the register set. When the limit value is exceeded an appropriateinterrupt is generated.

When the temperature value is applied to the input of 12 bit ADC 4910,the digitized temperature signal is applied to the input of the TEMPSFR/LIM register set 4930. As described previously, this value iscompared with a temperature limit value in the register set 4930, and ifthis value is exceeded, a TEMPIRQ is generated at output 4932. However,the output of the TEMP SFR/LIM register set 4930 is connected to theinput of an OR gate 4934. This is due to the fact that not enoughinterrupt resources are available for each of the SFR/LIM register set,so a limited number of the register sets have their outputs applied tothe input of OR gate 4934. The interrupt provided to the input of ORgate 4934 is also provided at the output 4936 of OR gate 4934. Thus,when the TEMP's IRQ is applied to input 4932, it will also be providedat the output pin 4936. When a digital value is applied to a particularSFR/LIM register set, the remaining SFR/LIM register sets are eachdisabled. Thus, when a digital signal associated with a particularregister set is being applied, that register set is the only registerset which is presently enabled.

Referring now to FIG. 50, there is more fully illustrated the process ofoperation of the SFR/LIM register sets. Initially, at step 5002 each ofthe VSENSE input voltage, the input voltage VIN and the temperature aremonitored by the above-described circuitry. When a particular SFR/LIMregister set determines at inquiry step 5004 that a limit value has beenexceeded, the interrupt is generated at step 5006. If inquiry step 5004determines that no value has been exceeded, control passes back to themonitoring step 5002. Once the interrupt 5006 has been generated andprovided to the controller 440 of the switched power supply, thecontroller will access at step 5008 the special function register set todetermine what the present problem may be.

Referring now to FIG. 51, there is illustrated a block diagram of thePLL block. The reference phase, i.e., an external or internallygenerated signal, is received on an input 5102 and input to one input ofa phase-frequency detector 5104. The output of this is input to a chargepump circuit 5106 which is operable to charge a node from a positivesourcing circuit or to discharge the node to a sinking circuit. This isconventional. The output of the charge pump circuit 5106 is input to aloop filter 5108 to generate a control voltage for a voltage controlledoscillator (VCO) 5110. This output is provided as the upper levelfrequency of, in this example, 400 MHz. This is input to one input of amultiplexer 5112. This is then output to a divide-by-two circuit 5114,which provides on the output a 200 MHz clock, this being the preferredDPWM clock. The output of block 5114 is then input to anotherdivide-by-two circuit 5116 to provide a 100 MHz clock signal, which istypically unused, which is then output to a third divide-by-two circuit5120, which provides a 50 MHz output and then to a divide-by-two block5122 to provide on the output thereof a 25 MHz signal for input to theother input of the phase-frequency detector 5104. This PLL provides the200 MHz clock for the DPWM clock. The filter clock is provided bydividing this by a factor of 20.

In an alternate operation, there is a test mode provided wherein theinput 5102 is input to the other input of a multiplexer 5112 forbypassing the PLL operation in the blocks 5104-5110. This allows thedividers to be directly controlled and the frequency of operations to becontrolled also.

Although the preferred embodiment has been described in detail, itshould be understood that various changes, substitutions and alterationscan be made therein without departing from the scope of the invention asdefined by the appended claims.

1. A system for controlling a soft start for a switching converter,comprising: a switched power supply for generating an output voltagesignal responsive to a compensation signal; a first analog to digitalconverter for comparing the output voltage signal to a correctedreference voltage signal and generating a voltage error signalresponsive thereto; a compensation circuit for generating drive signalsfor the switched power supply responsive to the voltage error signal; asecond analog to digital converter for converting the output voltagesignal into a digital voltage signal; a microcontroller for low passfiltering the digital voltage signal and for taking a derivative of thefiltered digital voltage signal to generate a voltage error offsetsignal; an offset circuit having a first input connected to receive thevoltage error offset signal and a second input connected to receive avoltage reference signal and an output providing the corrected referencevoltage signal, the offset circuit injecting the voltage error offsetsignal into the voltage reference signal to generate the correctedreference signal enabling the output voltage signal to operate linearlyover an entire voltage range of the output voltage signal.
 2. The systemof claim 1, wherein the offset circuit further includes a digital toanalog converter for generating the corrected reference voltageresponsive to the derivative signal and the voltage reference signal. 3.The system of claim 1, wherein the first analog to digital converterprovides a linear voltage response over a first voltage range.
 4. Thesystem of claim 3, wherein the voltage error offset signal alters thereference voltage to enable the second analog to digital converter andthe microcontroller to provide a linear voltage response over a secondvoltage range.
 5. The system of claim 1, wherein the first analog todigital converter comprises a flash analog to digital converter.
 6. Thesystem of claim 1, wherein the second analog to digital convertercomprises a 12 bit SAR analog to digital converter.
 7. A digitalcontroller for controlling the operation of a DC-DC switching converter,comprising: a digital feedback control system for receiving an analogvoltage representing an output of the switching converter and digitallyprocessing the analog voltage by comparing it to a corrected referencevoltage signal to generate a voltage error signal and generating drivesignals to control the operation of the switching converter to provide aregulated output voltage; a microcontroller for low pass filtering adigital voltage signal of the regulated output voltage and for taking aderivative of the filtered digital voltage signal to generate a voltageerror offset signal; an offset circuit having a first input connected toreceive the derivative signal and a second input connected to receive avoltage reference signal and an output providing the corrected referencevoltage signal, the offset circuit injecting the voltage error offsetsignal into the voltage reference signal to generate the correctedreference signal enabling the output voltage signal to operate linearlyover an entire voltage range of the output voltage signal.
 8. Thedigital controller of claim 7, wherein the digital controller furtherincludes: a first analog to digital converter for comparing theregulated output voltage to the corrected reference voltage signal andgenerating the voltage error signal responsive thereto; and a secondanalog to digital converter within the microcontroller for convertingthe regulated output voltage into the digital voltage signal.
 9. Thedigital controller of claim 7, wherein the offset circuit furtherincludes a digital to analog converter for generating the correctedreference voltage responsive to the derivative signal and the voltagereference signal.
 10. The digital controller of claim 8, wherein thefirst analog to digital converter provides a linear voltage responseover a first voltage range.
 11. The digital controller of claim 10,wherein the voltage error offset signal alters the reference voltage toenable the second analog to digital converter and the microcontroller toprovide a linear voltage response over a second voltage range.
 12. Asystem for controlling a soft start for a switching converter,comprising: a digital feedback control system for receiving an analogvoltage representing the output of the switching converter and digitallyprocessing the analog voltage by comparing it to a corrected referencevoltage signal to generate a voltage error signal and generating drivesignals to control the operation of the switching converter to provide aregulated output voltage; a microcontroller for providing a secondaryfeedback loop for generating the corrected reference voltage signal; andan offset circuit having a first input connected to receive thederivative signal and a second input connected to receive a voltagereference signal and an output providing the corrected reference voltagesignal, the offset circuit injecting the voltage error offset signalinto the voltage reference signal to generate the corrected referencesignal enabling the output voltage signal to operate linearly over anentire voltage range of the output voltage signal.
 13. The system ofclaim 12, wherein the digital controller further includes: a firstanalog to digital converter for comparing the regulated output voltagesignal to a corrected reference voltage signal and generating a voltageerror signal responsive thereto; and a second analog to digitalconverter within the microcontroller for converting the regulated outputvoltage signal into a digital voltage signal.
 14. The system of claim13, wherein the microcontroller is further configured to low pass filterthe digital voltage signal and to take a derivative of the filtereddigital voltage signal to generate the voltage error offset signal. 15.The system of claim 14, wherein the adder circuit further includes adigital to analog converter for generating the corrected referencevoltage responsive to the voltage error offset signal and the voltagereference signal.
 16. The system of claim 13, wherein the first analogto digital converter provides the linear voltage response over a firstvoltage range.
 17. The system of claim 16, wherein the voltage erroroffset signal alters the reference voltage to enable the second analogto digital converter and the microcontroller to provide a linear voltageresponse over a second voltage range.
 18. The system of claim 13,wherein the first analog to digital converter comprises a flash analogto digital converter.
 19. The system of claim 13, wherein the secondanalog to digital converter comprises a 12 bit SAR analog to digitalconverter.